Current mirror amplifier

Abstract

A current mirror amplifier in which the potentials appearing across the principal conduction paths of its master and slave transistors are maintained substantially the same. This is done by a differential-input, single-ended output amplifier, which is connected as a non-inverting amplifier in the direct-coupled feedback connection that conditions the master transistor to conduct applied input current via its principal conduction path.

Description

The present invention relates to transistor amplifiers and more particulary to current mirror amplifiers.

A current mirror amplifier is defined in connection with the present invention as an amplifier having a current gain substantially equal to the transconductance of a slave transistor divided by the transconductance of a master transistor. Each transistor has first and second electrodes and a principal conduction path therebetween and has a control electrode, with the conductances of the principal conduction path controlled in response to potential applied between the first and control electrodes. The master transistor has an input current applied to its principal conduction path and is provided with direct-coupled feedback between its second and control electrodes to apply a potential between its first and control electrodes that conditions its principal conduction path to conduct all or substantially all the input current. The potential between the first and control electrodes of the master transistor is applied between the first and control electrodes of the slave transistor to condition its principal conduction path to conduct the desired level of output current.

A basic reason for using a current mirror amplifier is that its current gain being determined by the ratio between the transconductances of its slave and master transistors, and the transconductances of transistor devices relying directly on certain physical dimensions that are readily proportioned as between transistors, the current gain of the current mirror amplifier can be accurately predicted despite shared variations in the transistors. Ideally, this current gain is constant despite expected changes in input and output current, potential levels and shared temperature variations.

Field effect transistors (FET's) can be used as the master and slave transistors. The first and second electrodes of an FET correspond to its source and drain electrodes; its principal conduction path is its channel; and its control electrode is its gate electrode. However, the transconductance of a FET changes, not only as a function of its physical dimensions, but also as a function of its source-to-drain potential. This dependence of FET transconductance upon source-to-drain potential is a substantially stronger second order effect than the dependence of the transconductance of a bipolar transistor on its emitter-to-collector potential; both effects are often referred to as "Early effect." Current mirror amplifier configurations in which the potentials appearing across the principal conduction paths of the master and slave transistors are not constrained to be equal, have undesirable inaccuracies in current gain when these transistors are FET's.

The present invention is embodied in a current mirror amplifier in which the potentials appearing across the principal conduction paths of its master and slave transistors are maintained substantially the same by a differential-input single-ended-output amplifier, which is connected as a non-inverting amplifier in the direct-coupled feedback connection of the master transistor.

In the drawing:

FIG. 1 is a schematic diagram of a complementary-conductivity FET amplifier, which includes for balanced-to-single-ended signal conversion purposes a current mirror amplifier embodying the present invention; and

FIG. 2 is a schematic diagram of a similar amplifier constructed with P-channel FET's and NPN bipolar transistors, which amplifier also includes a current mirror amplifier embodying the present invention.

FIG. 1 shows a long-tailed pair configuration comprising transistors Q₁ and Q₂ and a constant current generator S₁ that demands a current I₁ +I₂. Q₁ and Q₂ supply drain currents I₁ and I₂ that exhibit variations that are balanced with respect to each other, responsive to differential-mode potential applied between the non-inverting and inverting input terminals T₁ and T₂ connected to their respective gate electrodes. A current mirror amplifier CMA has an input terminal T₃ connected to the drain electrode of Q₁, has an output terminal T₄ connected to the source node N as the drain electrode of Q₂, and has a common terminal T₅. Current mirror amplifier CMA functions as a balanced-to-single-ended signal converter, inverting the differential-mode signal variations in the drain current I₁ of Q₁ for application to node N where they combine constructively with the different-mode signal variations in the drain current I₂ of Q₂. To satisfy Kirchoff's Law of Currents, the combined differential-mode current components flow through output load OL, developing a signal voltage thereacross in accordance with Ohm's Law.

Common-mode direct current components of the drain current of Q₂ and the output current of CMA combine destructively at node N, satisfying Kirchoff's Law of Currents, so that no direct current responsive to these common-mode components flows through output load OL connected between node N and the positive terminal of voltage supply S₂. So the direct component of potential at node N--that is, the quiescent potential--is the same as that at the positive terminal of supply S₂, connected at its negative terminal to reference ground, assuming that the direct component present in the applied signal is zero.

A further voltage supply S₃ is connected at its negative terminal to the positive terminal of voltage source S₂, and the positive terminal S₃ has the common terminal T₅ of current mirror amplifier CMA connected to it.

Q₃ and Q₄ are the master and slave transistors, respectively, in the CMA. The drain electrodes of Q₃ and Q₄ are connected to the input terminal T₃ of CMA and to the output terminal T₄ of CMA, respectively, and their source electrodes are connected to the common terminal T₅ of the CMA. Direct-coupled drain-to-gate feedback is applied to Q₃ by a differential-input, single-ended-output amplifier DA. Gate-to-gate connection provides Q₃ and Q₄ with like gate potentials. As with any feedback system, some capacitance such as C shown in dotted outline may be needed to augment the stray capacitance inherent in the current mirror amplifier CMA, to assure stability against self-oscillatory tendencies.

DA comprises: (a) FET's Q₅ and Q₆ connected in long-tailed pair configuration with constant current generator S₄, which demands a current I₅ +I₆ ; (b) FET's Q₇ and Q₈ connected in a subsidiary current mirror amplifier configuration SCMA; and (c) connection to the subsidiary current mirror amplifier SCMA as a balanced-to-signal-ended signal converter to convert the balanced collector currents I₅ and I₆ and Q₅ and Q₆ to single-ended form for application to the gate electrode of Q₃. More particularly, the current mirror amplifier SCMA has an input connection at the interconnected drain electrode of Q₇ and gate electrodes of Q₇ and Q₈, an output connection at the drain electrode of Q₈, and a common connection at the interconnection of the source electrodes of Q₇ and Q₈.

A current I₁ is withdrawn from input terminal T₃ of current mirror amplifier CMA. If I₁ exceeds the drain current I₃ of Q₃ there is a tendency for the gate electrode of Q₅ to be drawn to a potential less positive than the potential at the gate of Q₆. This reduces the conduction of Q₅ vis-a-vis Q₆ in their long-tailed pair connection, causing the portions of the current demand imposed by constant current generator S₄ that are satisfied by the source currents of Q₅ and Q₆ to be respectively relatively small and relatively large. This causes the drain current I₅ of Q₅ to be correspondingly small, which current is demanded from the input connection of subsidiary current mirror amplifier SCMA to cause a correspondingly small drain current to be supplied by Q₈. The drain current I₆ demanded by Q₆ being of the same amplitude as the source current of Q₆ is relatively large compared to the drain current supplied by Q₈. So the potential at the gate electrodes of Q₃ and Q₄, to which the drain electrodes of Q₆ and Q₈ connect, is drawn to a less positive potential. This increases the amplitude of the source-to-gate potentials of Q₃ and Q₄, increasing the conduction of Q₃ and Q₄ to adjust the drain current I₃ of Q₃ to equal I₁.

On the other hand, if the I₃ exceeds I₁, there is a tendency for the gate electrode of Q₅ to be drawn to a potential more positive than the potential at the gate electrode of Q₆. This increases the conduction of Q₅ vis-a-vis Q₆, causing I₅ to be relatively large compared to I₆. The relatively large I₅ withdrawn from the input connection of SCMA causes the collector current of Q₈ to be correspondingly large such that it exceeds I₆ in amplitude to draw the potential at the gate electrodes of Q₃ and Q₄ to more positive value. This reduces the amplitude of the source-to-gate potentials (V_GS 's) of Q₃ and Q₄, reducing the conduction of Q₃ and Q₄ to adjust I₃ to equal I₁.

If the transconductance of the amplifier DA is made sufficiently large, only a small difference between the gate potentials of Q₅ and Q₆ will be required to adjust the conduction of Q₃ to make I₃ equal to I₁. Ideally, of course, zero difference will be required. But in actuality, if Q₃ and Q₄ have appreciable gate-to-drain potentials, there will be some difference required to compensate for the mismatch in the transconductances (g_m 's) of Q₇ and Q₈ caused by their source-to-drain potentials (V_DS 's) differing somewhat. The V_DS of Q₇ will be its own V_GS. This error can be reduced at lower V_DS 's for Q₃ and Q₄, where a difference in their V_DS 's would cause the most difference between their g_m 's, by choosing I₅ +I₆ equal to I₁ +I₂ times the ratio of the g_m 's of Q₅ and Q₆ to the g_m 's of Q₃ and Q₄. At larger V_DS 's for Q₃ and Q₄ the small difference between their V_DS 's due to offset between the V_GS 's of Q₅ and Q₆ does not affect their relative g_m 's very much.

Since the g_m 's of Q₃ and Q₄ match well over the entire range, owing to amplifier DA maintaining their V_DS 's substantially equal, their similar V_GS 's will cause their source-to-drain currents to be in constant proportion. This is a 1:1 proportion where Q₃ and Q₄ are devices with matching dimensions, as would be used in a current mirror amplifier such as CMA used for balanced-to-single-ended conversion.

Modifications of the CMA where Q₃ and Q₄ have transconductances that are in ratio other than 1:1 are possible. Assuming Q₃ and Q₄ to be monolithic FET's, one can scale their g_m 's by making them with channels having differing width-to-length ratios, as is well known. Q₇ and Q₈ have g_m 's in the same ratio as those of Q₅ Q₆.

FIG. 2 shows a modification CMA' of the FIG. 1 current mirror amplifier CMA suitable for use in a monolithic integrated circuit technology in which NPN bipolar transistors and P-channel FET's are available. FET's Q₅ and Q₆ are replaced by the relatively high transconductance bipolar transistors. Only a few millivolts difference will appear between the base potentials of Q₅ ' and Q₆ ' despite one being substantially more conductive than the other. So error in the current gain of CMA' due to mismatch of the V_DS 's of Q₃ and Q₄ will be virtually non-existant. Some error due to the base currents of Q₅ and Q₆ will appear in the current gain of CMA', but it will be small so long as the current I₅ +I₆ is chosen so as not to be much larger than I₁ +I₂.

A number of modifications of the FIG. 1 and FIG. 2 will, in light of this application, suggest themselves to one skilled in the art of integrated circuit design. For example, one may alternatively replace Q₅ and Q₆ with FET devices having their transconductances multiplied by bipolar transistors. Q₃ and Q₄ may be provided with source degeneration resistors. Current generators S₁ and S₄ may be provided by simple resistive connections or from the collector or drain electrodes of fixed-bias transistors. A current mirror amplifier configuration according to the present invention is useful when all the transistors are of bipolar type, including Q₃ and Q₄, although the need for avoiding Early effect is less acute with bipolar transistors. Diodes may be introduced into the amplifier DA to adjust the V_DS 's of Q₇ and Q₈ to more equal value, if the conditions under which the current mirror amplifier is to operate are well defined. All such modifications which are in the spirit of the invention are to be considered within the scope of the following claims.

In the following claims a "transistor" is any current amplifying arrangement having input, common and output electrodes. The term "input" electrode is generic to the base electrode of a bipolar transistor and to the gate electrode of an FET; "common" electrode, to the emitter electrode of a bipolar transistor and to the source electrode of an FET; and "output" electrode to the collector electrode of a bipolar transistor and to the drain electrode of an FET, for example.

Claims

1. A current mirror amplifier comprising:

an input terminal;

an output terminal;

a common terminal;

first and second transistors of a first conductivity type, having respective input electrodes, having respective common electrodes connected to said common terminal, and having respective output electrodes connected respectively to said input terminal and to said output terminal; and

a differential-input single-ended-output amplifier having non-inverting and inverting input connections respectively at said input terminal and at said output terminal and having an output connection connected to an interconnection between the input electrodes of said first and said second transistors, said amplifier applying a signal to said interconnection having positive polarity when the potential at said non-inverting input connection is positive relative to the potential at said inverting input connection and having negative polarity when the potential at said non-inverting input connection is negative relative to the potential at said inverting input connection.

2. A current mirror amplifier as set forth in claim 1 wherein said differential-input single-ended-output amplifier includes:

third and fourth transistors of a second conductivity type complementary to said first conductivity type, having respective input electrodes to which said input terminal and said output terminal are respectively connected, having respective common electrodes, and having respective output electrodes;

a subsidiary current mirror amplifier having an input connection to which the output electrode of said third transistor connects, having an output connection to which the output electrode of said fourth transistor is connected, and having a common connection;

means for applying quiescent current between the common terminal of said subsidiary current mirror amplifier and an interconnection between the common electrodes of said third and fourth transistors; and

means connecting the output connection of said subsidiary current mirror amplifier to said interconnection between the input electrodes of said first and said second transistors.

3. A current mirror amplifier as set forth in claim 2 wherein the common connection of said subsidiary current mirror amplifier is at said common terminal.

4. A circuit for maintaining the potential at a first node substantially equal to the potential at a second node, said circuit comprising:

first and second supply terminals for receiving an operating potential therebetween;

current conductive impedance means connecting said first node to said first supply terminal;

first, second and third transistors, each having first and second electrodes and a controlled conductivity principal current conduction path therebetween and having a third or control electrode, the conductivity of its principal current conduction path being controlled responsive to potential appearing between its first and third electrodes, said first and second transistors being of the same conductivity type as each other,

means connecting said first and second transistors in long-tailed-pair configuration including an interconnection of their respective first electrodes, means for applying current to said interconnection of said respective first electrodes, a connection of the third electrode of said first transistor to said second node, and a connection of the third electrode of said second transistor to said first node,

a third node to which the second electrode of said first transistor connects;

means connecting said third transistor in common-first-electrode amplifier configuration including

means connecting the first electrode of said third transistor to one of said first and second supply terminals,

means direct coupling said third node to the third electrode of said third transistor, and including

a connection of the second electrode of said third transistor to said first node; and

current amplifier means having an input connection from the second electrode of said second transistor and having an output connection to said third node for differentially combining currents from the second electrodes of said first and second transistors to a single-ended current.

5. A combination as set forth in claim 4 wherein said first and second transistors are so proportioned respective to each other as to have the conductivities of their respective principal current conduction paths be in 1 :G ratio with each other for similar first-to-third-electrode potentials and wherein the current gain of said current amplifier is -G, G being a positive number.

6. A combination as set forth in claim 4 wherein said third transistor is of opposite conductivity type from that of said first and second transistor, and wherein said means connecting the first electrode of said third transistor to one of said first and second supply terminals consists of

means connecting the first electrode of said third transistor to said first supply terminal.