A series resonant inverter is controlled to provide a substantially constant output voltage to a load. The control utilizes a combination of optimal control methods and phase modulation to enable time optimal responses to changes in state of the system. State determinants (including resonant capacitor voltage, resonant inductor current, source voltage, and output load voltage) are continuously monitored, and an optimal control signal is generated therefrom. When operating within the operable frequency range of the inverter's controllable switch means, frequency is varied to maintain proper operation. When operating at an extremity of the operable frequency range, phase modulation is employed.
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9. A method for controlling a resonant inverter, said inverter having controllable switch means for producing a rectangular wave signal when coupled to an external dc supply and applying said signal to a series resonant circuit which comprises a capacitor and an inductor, the output of said resonant inverter providing a substantially constant output voltage to a load, said control method comprising the steps of:
continuously monitoring state determinants comprising voltage across said capacitor, current through said inductor, said rectangular wave signal the dc supply voltage, and said output voltage; generating an optimal control signal corresponding to a predetermined combination of the instantaneous values of said state determinants; frequency modulating said rectangular wave signal applied to said series resonant circuit so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is within the operable frequency range thereof; and generating a phase modulation angle signal for phase modulating said rectangular wave signal and modifying said optimal control signal in accordance therewith so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is at an extremity of the operable frequency range thereof.
5. At An improved control for a resonant inverter, said inverter including a series resonant circuit which comprises a capacitor and an inductor, said inverter further including controllable switch means for producing a rectangular are wave voltage when coupled to an external dc supply and applying said voltage to said series resonant circuit, the output of said resonant inverter providing a substantially constant output voltage to a load, said improved control comprising:
state determinant sensing means for continuously monitoring converter state determinants comprising voltage across said capacitor, current through said inductor, the rectangular wave voltage and applying said voltage applied to said series resonant circuit dc supply voltage, and the output voltage; optimal control means responsive to said state determinant sensing means for generating an optimal control signal corresponding to the instantaneous values of said state determinants; first control means responsive to said optimal control signal for controlling the output voltage by frequency modulating the rectangular wave voltage applied to said series resonant circuit so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is within the operable frequency range thereof; and second controls control means responsive to said optimal control signal for controlling the output voltage by providing a phase modulation angle signal for phase modulating the rectangular wave voltage applied to said series resonant circuit and modifying said optimal control signal in accordance therewith so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is at an extremity of the operable frequency range thereof.
1. An improved dc-to-dc converter, comprising:
a resonant inverter having two pairs of controllable switch means, the switch means of each pair being connected in series and each pair of the series-connected switch means being adapted to be connected in parallel across an external dc supply; a series resonant circuit connected between the junctions of said controllable switch means and comprising a capacitor and an inductor, said inverter being adapted to apply a rectangular wave voltage to said series resonant circuit; a full wave rectifier inductively coupled to said series resonant circuit, the output of said rectifier being adapted to supply a substantially constant preselected output voltage to a load; state determinant sensing means for continuously monitoring converter state determinants comprising voltage across said capacitor, current through said inductor, the rectangular wave voltage applied to said series resonant circuit dc supply voltage, and the output voltage; optimal control means responsive to said state determinant sensing means for generating an optimal control signal corresponding to the instantaneous values of said state determinants; first control means responsive to said optimal control signal for controlling the output voltage by frequency modulating the rectangular wave voltage applied to said series resonant circuit so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is within the operable frequency range thereof; and second control means responsive to said optimal control signal for controlling the output voltage by providing a phase modulation angle signal for phase modulating the rectangular wave voltage applied to said series resonant circuit and modifying said optimal control signal in accordance therewith so as to maintain stable operation of said series resonant circuit when the operating frequency of said controllable switch means is at an extremity of the operable frequency range thereof.
2. The improved converter of
frequency measuring means coupled to the output of said inverter for determining when the operating frequency of said controllable switch means is at an extremity of the operable range thereof.
3. The improved converter of
frequency modulation means for generating a frequency modulation signal; comparison means for comparing said frequency modulation signal with said optimal control signal and for generating a difference signal resulting therefrom; and frequency control means responsive to said difference signal for generating a frequency control signal for varying the operating frequency of said controllable switch means.
4. The improved converter of
sawtooth generator means responsive to said frequency control signal for generating a ramp voltage; second comparison means for comparing said ramp voltage with said phase modulation angle signal; and flip-flop means responsive to said frequency control signal and to the output signal of said second comparison means, said flip-flop means being coupled to said controllable switch means for providing control signals to vary the operating frequency of said controllable switch means when operating within the operable frequency range thereof and to phase modulate the rectangular wave voltage when operating at an extremity of the operable frequency range.
6. The improved control of
frequency measuring means coupled to the output of said inverter for determining when the operating frequency of said controllable switch means is at an extremity of the operable range thereof.
7. The improved control of
frequency modulation means for generating a frequency modulation signal; comparison means for comparing said frequency modulation signal with said optimal control signal and for generating a difference signal resulting therefrom; and frequency control means responsive to said differences difference signal for generating a frequency control signal for varying the operating frequency of said controllable switch means.
8. The improved control of
sawtooth generator means responsive to said frequency control signal for generating a ramp voltage; second comparison means for comparing said ramp voltage with said phase modulation angle signal; and flip-flop means responsive to said frequency control signal and to the output signal of said second comparison means, said flip-flop means being coupled to said controllable switch means for providing control signals to vary the operating frequency of said controllable switch means when operating within the operable frequency range thereof and to phase modulate the rectangular wave voltage when operating at an extremity of the operable frequency range.
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The present invention relates generally to resonant inverters. More particularly, this invention relates to a series resonant inverter with improved control which utilizes a method of optimal control in combination with phase modulation to maintain substantially constant output voltage over a wide range of operating conditions.
Resonant inverters advantageously have low switching losses and low switching stresses. However, resonant operation is complex due to the fast dynamics of the high-frequency resonant tank circuit; and, hence, control is difficult. Disadvantageously, when input power or output load conditions vary, output voltage or current control cannot be achieved through the use of usual control techniques. For example, one known resonant inverter output load voltage or current control method is to vary the frequency of the rectangular wave signal supplied to the resonant circuit by the inverter via closed loop control. Commonly assigned U.S. Pat. No. 4,541,041, issued on Sept. 10, 1985 to J. N. Park and R. L. Steigerwald, which is hereby incorporated by reference, discloses in part such a frequency control technique. Briefly explained, the resonant nature of the circuit allows for control of output voltage or current through variation of the frequency at which the inverter's controllable switch means operate. Such a frequency control method has been formed satisfactory under normal output load conditions for particular types of resonant inverters (i.e., heavy or medium load conditions for a series resonant inverter and light load conditions for a parallel resonant inverter). The drawback to frequency control, however, is that it may be inadequate to maintain a desired output voltage or current under extended output load conditions (i.e., light or no load conditions for a series resonant inverter and heavy load conditions for a parallel resonant inverter).
In particular, frequency control of a series resonant inverter will normally be adequate to maintain a desired output voltage during heavy or medium load conditions (i.e., low load resistance); that is, for heavy or medium load conditions, a series resonant circuit has a high quality factor Q and thus a good dynamic range of voltage or current change as frequency is varied. However, under extended or light output load conditions (i.e., high load resistance) the series resonant circuit exhibits a low quality factor Q and thus only a small dynamic range of output voltage or current change can be achieved as a function of frequency. As a result, for a series resonant inverter, it may be impossible to maintain a desired output voltage or current under light load and no load conditions solely with frequency control.
A resonant inverter control which provides an improved dynamic range of output voltage or current control is disclosed in U.S. Pat. No. 4,672,528, issued June 9, 1987 to J. N. Park and R. L. Steigerwald and assigned to the assignee of the present invention. This patent, which is hereby incorporated by reference, describes a resonant inverter which is controlled using either a frequency control mode or a phase shift control mode. In the frequency control mode, output voltage is controlled by varying the frequency of the rectangular wave signal supplied to the resonant circuit within an operable range of the controllable switch means. Selecting means allows the control to operate in the phase shift control mode when the frequency of the rectangular wave signal is at an extremity of the operable range of the controllable switch means.
Another method of resonant inverter control, which is derived from optimal control theory and state plane analysis, is presented in "Resonant Power Processors: Part II-Methods of Control" by Ramesh Oruganti and Fred C. Lee, 1984 Industry Applications Society Proceedings, pp. 868-878, and is hereby incorporated by reference. According to this method, hereinafter designated "optimal trajectory control" to be described in detail below, each state trajectory corresponds to particular values of instantaneous resonant tank energy, output voltage, output current and switching frequency. These state trajectories are utilized to define a control law for the inverter control system which enables a series resonant inverter to respond quickly to load and control requirements. Disadvantageously, however, in the method of "optimal trajectory control", as it presently exists, the controlled range of output voltages is limited in the same manner as the hereinabove described conventional frequency control method.
Accordingly, it is an object of the present invention to provide a new and improved resonant inverter exhibiting an improved dynamic range of output load voltage control.
Another object of this invention is to provide a new and improved resonant inverter control which utilizes a combination of optimal control methods and phase modulation to maintain output load voltage substantially constant during all loading conditions.
Still another object of this invention is to provide a new and improved resonant inverter control which switches automatically between different control means to maintain a substantially constant output load voltage.
Yet another object of the present invention is to provide an improved method of controlling a resonant inverter in order to maintain a desired output load voltage.
In accordance with the present invention, a new and improved resonant inverter is controlled using a combination of optimal trajectory control and phase modulation. In particular, optimal control means are employed to continuously monitor resonant capacitor voltage, resonant inductor current, voltage applied to the resonant tank circuitk3 (vc -FVo) k3 (vc +FVo) is added to the aforementioned signal Fk3 Vs cos φ by summer 42 to yield the signal k3 (vc -FVo -FVs cos φ). k3 (vc +FVo +FVs cosφ). The latter signal is inputted to a multiplier 44 which performs a squaring operation. The resulting squared signal k2 (vc -FVo -FVs cos φ)2 k2 (vc +FVo +FVs cos φ)2 is added to the hereinabove derived signal k2 (Zo iL)2 by a summer 46 and, as shown in FIG. 7b, is then inputted to gain amplifier 48 having the transfer function -k4 /k2 where k4 is a constant. The output of amplifier 48 is a signal -k4 [(vc -FVo -FVs cos φ)2 +(Zo iL)2 ] -k4 [(vc +FVo +FVs cos φ)2 +(Zo iL)2 ], which is hereinafter referred to as the optimal control signal.
Control signal VCONTROL is provided to a frequency modulation controller 50 and a phase modulation controller 52. The transfer function of frequency modulation controller 50 is shown in FIG. 7b and may be represented mathematically as: ##EQU3## where VF is the output voltage of frequency modulation controller 50, VT is a threshold voltage representing operation at an extremity of the operable frequency range for the controllable switch means, and C1 is a constant. Voltage VF is added in a summer 54 to the output signal of gain amplifier 48, and the result is inputted to the non-inverting input of a comparator 56. The output signal from comparator 56 is supplied to a saw-tooth generator 58.
The transfer function of phase modulation controller 52 is also shown in FIG. 7b and may be represented mathematically as: ##EQU4## where Vφ is the output voltage from phase modulation controller 52, Vφ being proportional to phase modulation angle φ, and C2 is a constant. Voltage Vφ is inputted to the inverting input of a comparator 60. The output signal VG of sawtooth generator 58 is supplied to the noninverting input of comparator 60. Voltage Vφ is also supplied to multiplier 31 for which cos φ is the multiplicative factor.
The output signals CP1 and CP2 from comparators 56 and 60, respectively, provide the clock pulses for D-type (delay) flip-flops 62 and 64, respectively. As will be appreciated by those of skill in the art, since the signal at output D flip-flop 62 is supplied to the D1 input of D flip-flop 62, D flip-flop 62 is a divide-by-two flip-flop; that is, the output frequency is one-half that of the clock frequency. The output signals from the D flip-flops control the base drive circuitry 65a-65d for the respective switching devices S1-S4. Suitable base drive circuitry is well-known in the art.
In operation, since the output signal from comparator 56 which provides clock pulses to the divide-by-two D flip-flop 62 also drives sawtooth generator 58, the sawtooth generator produces a voltage ramp signal VG operating at twice the frequency of gate drive circuitry 65a-65d. In particular, the voltage ramp signal VG resets to zero each time the output signal at Q1 of D flip-flop 62 transitions from logic level 0 to 1 or 1 to 0. The output ramp voltage of sawtooth generator 58 is compared with voltage Vφ by comparator 60 which provides clock pulses for D flip-flop 64. For a positive edge triggered D flip-flop 64, for example, when the output signal of comparator 60 transitions from a low logic level to a high logic level, the signal at output Q2 of D-flip-flop 64 latches to the same value as the signal at output Q1 of D flip-flop 62.
For VCONTROL <VT, the output voltage VF of frequency modulation controller 50 is C1 VCONTROL, and the output voltage Vφ of phase modulation controller 52 is zero, thus indicating that phase modulation angle φ=0. Therefore, since the value of phase modulation angle φ is provided to multiplier 31, and cos φ=1 for φ=0, there is no phase modulation. On the other hand, there is frequency modulation. That is, the output voltage C1 VCONTROL of frequency modulation controller 50 is added to the output signal of summing amplifier 48 and applied to the non-inverting input of comparator 56. The output signal CP1 of comparator 56 provides clock pulses to D flip-flop 62 to toggle its state and, as stated above, also drives sawtooth generator 58. The output voltage Vc VG of the sawtooth generator is compared with voltage Vφ =0 by comparator 60 which provides clock pulses CP2 to D flip-flop 64. As a result, D flip-flop 64 is toggled almost simultaneously with D flip-flop 62. In this way, for VCONTROL <VT, frequency modulation using optimal control is achieved when operating within the operable frequency range of the switching devices.
For VCONTROL ≧VT, the output voltage VF of frequency modulation controller 50 is C1 VT, a constant, so that the switching frequency of switching devices S1, S2, S3 and S4 is fixed at an extremity of the operable frequency range thereof. Under these conditions, the output voltage Vφ of phase modulation controller 52 is C2 (VCONTROL -VT). This voltage Vφ is compared with the output signal VG of sawtooth generator 58 by comparator 60. As a result, the clock pulses CP2 from comparator 60 to D flip-flop 64 are delayed by an amount of time proportional to phase modulation angle φ. Voltage Vφ also enables multiplier 31 to multiple source voltage VS by cos φ. In this way, phase modulation is employed to produce the tri-level voltage waveform shown in FIG. 4B for controlling the series resonant inverter. By thus combining a method of optimal trajectory control with phase modulation, a broader dynamic range of output load voltage can be achieved under all operating conditions.
FIGS. 8a-8i are waveforms that illustrate in detail the operation of the new resonant inverter control for a specific case of VCONTROL >VT. For simplicity, assume the output signal CP1 of comparator 56 has a constant pulse width and is represented by the signal of FIG. 8a. For a positive edge-triggered D flip-flop 62, the output signals at Q1 and Q1 respectively, are illustrated in FIGS. 8b and 8c, respectively. Voltage ramp signal VG from sawtooth generator 58, which is reset each time the output signals from D flip-flop 62 change state, is shown in FIG. 8d. Voltage Vφ, which determines the phase modulation angle φ, is illustrated as a voltage between 0 and 10 V in FIG. 8e. For this example, voltage Vφ =5 V. The output signal CP2 of comparator 60, determined by comparing voltage Vφ with the output ramp voltage VG of sawtooth generator 58, is represented in FIG. 8f and constitutes clock pulses for D flip-flop 64. For a positive edge-triggered D flip-flop 64, the output signals at Q2 and Q2 Q2 respectively, are illustrated in FIGS. 8g and 8h, respectively. The flip-flop output signals at Q1, Q1 Q1, Q2 and Q2 Q2, respectively, control the base drive circuitry 65a-65 d, respectively, and produce as a result the tri-level phase modulated signal shown in FIG. 8i. From FIG. 8i and the equation for phase modulation angle φ given hereinabove, it can be seen that phase modulation angle φ=π/4 radians for this example.
While the preferred embodiments of the present invention have been shown and described herein, it will be obvious that such embodiments are provided by way of example only. Numerous variations, changes and substitutions will occur to those of skill in the art without departing from the invention herein. Accordingly, it is intended that the invention be limited only by the spirit and scope of the appended claims.
Park, John N., Kuo, Ming H., Schutten, Michael J.
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