Spread spectrum communications system

Abstract

A spread spectrum communication system for direct sequence transmission of digital information having a modulation format which is particularly suitable for indoor communication within residential, office and industrial structures. The modulation format combines BPSK or MSK spreading with FM carrier modulation by data bits and a carrier frequency shift whose magnitude is related to both a chip rate and a spreading sequence length. The carrier, chip clock and data clock are all synchronous and the sequence length is an integral submultiple of the bit length. The system reduces the frequency error between the transmitter chip clock and the receiver chip clock to permit the elimination of a code phase tracking loop in the receiver to reduce the receiver complexity. The receiver has an extended dynamic range which makes possible the reception of very strong signal without an automatic gain control loop (AGC) as well as reducing the time needed for code phase acquisition. The transmission system is highly resistant to CW jamming and short distance multipath effects.

Description

The present invention relates to a method and apparatus for the transmission of digital data in a spread spectrum communications system. In particular, it relates to a spread spectrum communication system for use in transmitting alarm data within a building or other structure which can contain sources of electromagnetic interference and in which there is a potential for short distance multipath cancellation effects.

Techniques for achieving direct sequence spread spectrum modulation are well known. The most commonly accepted method involves generating a periodical, relatively high frequency, repetitive pseudo-noise code (PN code) and effectively mixing the data signal with this PN code using exclusive OR gates or a balanced mixer. The resulting signal is characterized by very wide bandwidth and very low spectral density. The direct sequence spread spectrum modulation is particularly attractive for data transmission within a building (compared to narrow band radio frequency carriers) in that the low spectral density characteristic of such signals reduces the tendency for interference with other radio sensitive equipment. Also, spread spectrum techniques have the intrinsic ability to reduce interference between multiply reflected versions of the transmitted signal, since reflections which differ in phase only slightly from the initial signal being decoded result in low signal correlation and consequent suppression of such signals. In a similar manner, the same spread spectrum techniques can be used to suppress other jamming signals which may be present in the medium and interfere with the received signal.

It is further well known that communications within buildings or other structures can cause substantial attenuation of the original signal. Consequently, a receiver for such systems must have a wide dynamic range, in order to be able to acquire weak signals as well as strong signals, particularly if the communication is to be performed at random between a plurality of transmitters and receivers.

Both the jamming and multipath phenomena for direct sequence spread spectrum techniques have been discussed in papers and textbooks and patent documents. However, most of these have restricted the scope of their attention to cases where the jamming signals are introduced deliberately into the medium and intelligently controlled to cause the most harm. This results in the objective being to determine the method most immune to the interference under worst case conditions. It is has also usually been tacitly assumed that the data mixing with the PN code and modulating the radio frequency carrier should result in a modulated waveform which will not permit demodulation of the data without prior synchronization and correlation to the same PN code used for the original modulation. This results in some restriction being placed on the way the data, PN code and carrier are combined to convey the information, as well as restrictions being placed on the relationship between the data rate and the sequence repetition rate.

The multipath delay is usually considered to be relatively long. That is, the delay between the original data and the delayed replica is assumed to be much longer than one chip time. The mechanisms devised to deal with this multipath problem are different than those needed to suppress so called "selective fading" caused by electromagnetic field cancellation which occurs when the original signal and the delayed replica arrive at the receiver in opposite phase. This occurs in the case where the delay is approximately (2*k+1)*wv1/2, where k is an integer and wvl is the carrier wavelength where both the original and the replica have approximately the same amplitude. However, it has been suggested in the scientific literature that short delays are one of the most important factors to consider for indoor communications.

The above-delineated restrictions are a natural consequence of the major original applications of spread spectrum communications, that is, military applications in generally hostile communications environment in which jammers are intelligent and active and communication is attempted over distances much larger than building interiors. Furthermore, in these cases, a major concern is a need to hide or at least provide a low probability of detection of the attempted communication by enemy eavesdroppers.

Only recently has the Federal Communications Commission (FCC) permitted the utilization of spread spectrum communications techniques for commercial applications. The FCC has now permitted such operations in three separate frequency bands. Thus, there is a need for a spread spectrum communications system which addresses the needs of commercial applications and which are not restricted by the needs of military applications.

SUMMARY OF THE INVENTION

It is the general object of the present invention to provide an improved, method and apparatus for transmitting spread spectrum signals.

Another object of the invention is to provide a method and apparatus for transmitting spread spectrum signals which has jamming rejection substantially better than expected for a given spreading sequence length.

A further object of the invention is to provide a method and apparatus for transmitting spread spectrum signals in which short distance multipath cancellation of the type encountered in residential and office buildings is eliminated or substantially reduced.

Yet another object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals in which there is a reduction of the frequency error between the transmitter chip clock and the receiver chip clock, which permits the elimination of a code phase tracking loop in the receiver, which in turn substantially reduces receiver complexity.

A further object of the present invention is to provide a method and apparatus for transmission of spread spectrum signals in which there is an extension of the dynamic range of the receiver which makes possible the reception of very strong signals without an automatic gain control loop (AGC) while at the same time reducing the time needed for code phase acquisition.

Another object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals that provide the receiver with immunity from interference from CW signals generated by personal computer crystal oscillators or the like.

Another object of the invention is to provide a method and apparatus for transmitting spread spectrum signals in which data, chip clock and carrier are synchronous.

A further object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals in which the bit period length (Tb) is N (where N is an integer greater than 1) times longer than the pseudo noise (PN) sequence period (Ts) which results in a modified spectrum of the spread signal and consequently the spectrum of the jammer being spread by the receiver's despreading mixture.

Yet another object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals in which the transmitter center frequency is frequency modulated at a data rate with a low deviation and in addition is frequency shifted at a lower rate by a large increment relative to the chip rate and sequence length and in which the same data packet consisting of a preamble and data is repeated after each frequency shift.

A further object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals in which the data packet consists of a preamble and data following the preamble, the preamble being longer than the data and comprising a period in which the carrier is not modulated and the data comprising a period in which data bits modulate the carrier center frequency.

A still further object of the present invention is to provide a method and apparatus for transmitting spread spectrum signals in which the modulated carrier is then spread to the desired transmission bandwidth by applying a PN code sequence to a balanced mixer or other device to obtain BPSK spreading.

Yet another object of the invention is to provide a receiver for receiving a spread spectrum signal in which the resultant IF spectral components of a CW jamming waveform present in the RF section of the receiver are separated in frequency by an amount larger than the bandwidth occupied by the desired despread signal.

A still further object of the present invention is to provide a method and apparatus for transmitting a spread spectrum signal in which a single CW jammer can produce at most only one spectral component of the spread jammer present in the passband of the narrow band FM receiver following the despreading mixer.

These and other objects, advantages and features are achieved by a method of transmitting digital data. A high frequency carrier is generated at a first carrier frequency. The frequency of said carrier is modulated by data bits assembled in data packets of predetermined format. Generating a high frequency carrier at said first carrier frequency and shifting the nominal frequency of said carrier by a frequency increment DF to obtain a second carrier frequency, where DF=K * 1/Ts+dF, dF=approximately 1/(2*Ts), K is an integer, and Ts is a pseudo noise (PN) sequence period. Modulating the frequency of said second carrier frequency by data bits assembled in data packets of said predetermined format. The carrier is spread at said first and said second frequency, during a transmission time, by a predetermined PN sequence having a period Ts related to a bit time Tb, where Tb=N * Ts, N being an integer >1. This generates a spread carrier spectrum comprising many spectral components separated by 1/Ts in which the amplitudes of said components are reduced by the spreading function. The resulting components are modulated in the same manner as said carrier prior to spreading. The resulting modulated component spectra do not overlap, so that components and the modulation imposed upon them are separable from other components by a narrow band filter.

Another aspect of the invention includes a transmitter for transmitting digital data. A carrier frequency generator means generates a carrier frequency signal at a first frequency. A modulator means modulates said carrier by data bits. A frequency shift means generates a second carrier frequency offset from said first carrier frequency by DF, where DF=K * 1/Ts+dF, dF=approximately 1/(2*Ts), K is an integer, and Ts is a pseudo noise (PN) sequence period. A pseudo random sequence generator means generates a predetermined PN sequence. A carrier spread means spreads power of said first and second carrier frequencies by said pseudo noise sequence.

A further aspect of the invention includes a receiver for a spread spectrum signal. A means for receiving a spread carrier radio signal generates a spread carrier electrical signal. A means generates a predetermined pseudo noise sequence (PN sequence)combined with a local oscillator signal. A multiplying means coupled to said receiving means and to said generating means multiplies said spread carrier electrical signal by said predetermined PN sequence for collapsing the bandwidth of the received spread carrier when the local PN sequence phase is in agreement with the sequence phase imposed on the received spread carrier by a transmitter means. This spreads any jamming signals which are received along with the transmitted spread signal into many components separated by 1/Ts intervals and reduces their amplitude by the spreading function. A narrow band FM receiver means coupled to an output of said multiplying means recovers data. A means coupled to said receiving means alternately receives a radio signal on one of two preselected frequencies. A frequency switch means coupled to said means for alternately receiving radio signals switches the received frequency between a first and a second of said preselected frequencies at predetermined time intervals.

A still further aspect of the invention includes a system for transmitting and receiving digital data. A means generates a high frequency carrier at a first carrier frequency. A means modulates the frequency of said carrier by data bits assembled in data packets of predetermined format. A means shifts the nominal frequency of said carrier by a frequency increment DF to obtain a second carrier frequency, where DF=K * 1/Ts+dF, dF=approximately 1/(2*Ts), K is an integer, and Ts is a pseudo noise (PN) sequence period. A means spreads said carrier at said first and said second frequency, during a transmission time, by a predetermined PN sequence having a period Ts related to a bit time Tb where Tb N * Ts, N being an integer >1. This generates a spread carrier spectrum comprising many spectral components separated by 1/Ts. The amplitudes of the components are reduced by the spreading function. Each of the components are modulated in the same manner as said carrier prior to spreading, and do not overlap. A means receives 6*L*Tb*Dc 6*Ls*Tb*Dc where L Ls is the sequence length, Tb is the bit time and Dc is a designer choice constant which is usually 1 or 2. The receiver dwell time Td on one frequency has in the preferred embodiment a time of 2*L*Tb*Dc2*Ls*Tb*Dc. The receiver frequency shift signal, the frequency of the reference oscillator and the frequency of the local oscillator are shown in FIG. 6. FIG. 6a represents the frequency shift signal generated by microprocessor 570 and input to summation amplifier 568 via line 524. The time Td is the receiver dwell time in the search mode, which has been described above. During time T' the reference oscillator output shown in FIG. 6b is at a frequency Fr. During a time T" the frequency of the reference oscillator is at a frequency Fr+df. The frequency of the local oscillator synthesizer is shown in FIG. 6c. During the time T' it is at frequency of Ftx1-Fif and during the period T" it is at a frequency Ftx2-Fif.

Upon PN code acquisition the signal on line 506 (FIG. 5) consists of the despread, narrow band FM modulated signal and potentially a spread jamming signal whose spectral components are spaced by 1/Ts, where Ts is the spreading sequence period. The exact placement of the spectral components of the spread jamming signal with respect to the desired signal is determined by the jamming frequency prior to spreading.

FIGS. 7a and 7b show the spectrum of the signal received at the receiver antenna during time T1 and T2 with a jamming signal present in the spectrum. The signal 702 in FIG. 7a is the signal transmitted during time T1. The signal 704 is the jammer at a power level of P_j and a frequency of F_j. The typical (sin x/x)² envelope produced by modulating rectangular pulses is shown by the attenuation between lobes 702, 706 and 708. The (sin x/x)² characteristic causes lobe 706 to be 13 dB lower than the desired signal lobe 702. Lobe 708 is lower than lobe 706 in accordance with the (sin x/x)² rolloff. In FIG. 7b, 710 represents the signal transmitted during time T2. Reference numeral 712 represents the jammer at a frequency F_j ' and a power P_j. Reference numerals 714 and 716 represent the (sin x/x)² envelope. Similarly to FIG. 7a, lobe 714 is 13 dB lower than lobe 710 and lobe 716 is lower than lobe 714 in accordance with the (sin x/x)² rolloff. FIGS. 7c and 7d illustrate the envelope of the spread jamming signal after conversion at the output of mixer 552 (FIG. 5) during time T1 and T2. As is well known to those skilled in the art, when a CW signal is passed through a mixer of a spread spectrum receiver, it is acted on similarly to the signal passing through multiplier 112 of the transmitter (FIG. 1). That is, the signal is spread by the PN sequence This produces the spectrum shown in FIGS. 7c-7f. In FIG. 7c, reference numeral 720 represents the jammer envelope after the mixer 552 during the time T'. Reference numeral 722 represents the peak power of the jammer signal in db as P_j '-10LogL and 724 represents the power P_j. In FIG. 7d, 730 represents the jammer envelope after the mixer 552 during the time T". Reference numeral 732 represents the power P_j " and reference numeral 734 is the power P_j -10LogL.

FIGS. 7e and 7f show the close-in spectrum of the spread jamming signal at the output of mixer 552 (FIG. 5) during time T1 and T2 at points 724 (P_j ') and 732 (P_j ") and also the relationship of the jamming signal component to the despread desired signal and the receiver IF bandwidth. The spectrum shown in FIGS. 7e and 7f is an expansion of the points in FIGS. 7c and 7d along line 728. In FIG. 7e, 740 represents the despread received signal and 742 represents the bandwidth of the IF band pass filter. Reference numeral 744 represents the jammer at power P_j ' and 746 and 748 represent the close-in spectrum jammer components during time T'. In FIG. 7f, reference numeral 750 represents the despread received signal. Reference numeral 752 represents the bandwidth of the IF band pass filter. Reference numeral 754 represents the-jammer signal at a power P_j " and reference numeral 756 and 758 represent the close-in spectrum of the jammer components during time T".

The jamming signal envelope, during the time T1, is centered around the frequency F_j -Flo' where Flo' is the local oscillator frequency-during time T1, which is above the Fif by approximately F_j -Ftx1, where F_j is the jammer frequency and Ftx1 is the center frequency of the signal transmitted at time T1. Similarly, the jamming signal envelope during the time T2 is centered around F_j -Flo" where Flo" is the local oscillator frequency at time T2, which is below Fif by approximately Ftx2-F_j, where Ftx2 is the center frequency of the signal transmitted at time T2. This is due to the frequency conversion in the mixer 552 based on changing the frequency of the local oscillator. Because of the (sin x/x)² shape of the envelope, the power level at P_j " (which is in the second lobe) is approximately 10 dB lower than the power level at P_j ', where P_j ' and P_j " are the jamming signal spectral component power level translated to the IF frequency during times T1 and T2, respectively.

The drawings illustrate that if the despread signal spectrum is narrow enough so that the receiver bandwidth is less than 1/(2*Ts) and the frequency difference between Ftx1 and Ftx2 is equal to Fr plus approximately 1/(2Ts) then at least one reception, that is, Ftx1 or Ftx2, is not corrupted by the jamming signal interference. This is because a narrow-bandwidth receiver filter is effectively between the spectral lines of the spectrum jamming signal. Furthermore, the jamming spectral lines are down by the spreading factor plus an additional factor related to the spread jamming signals (sin x/x)² spectral envelope. This can be seen in FIGS. 7e and 7f at P_j ' which is in the passband whereas P_j " is not.

As a result, the frequency shift technique is effective in suppressing the non-intentional CW jamming signals, such as those created by computers, terminals, and other devices employing crystal based or other frequency determining elements as part of clock timing circuits. Some gain is also obtained in the suppression of narrow band as well as wide band modulated jamming signals due to the spreading factor and result in the (sin x/x)² envelope of the spread jamming signal caused by the correlating mixer.

The frequency shift technique described above is also an effective means of suppressing short delay multipath signals frequently encountered when a high frequency carrier is used to communicate inside of buildings. The present invention provides good results for both long delay and, more importantly, for short delay multipath effects in a spread spectrum communication system in which the delays are much shorter than the chip time. Traditional methods are less effective on this type of multipath delay. FIG. 8 illustrates the short multipath effect on the spread spectrum signal. FIG. 8a represents a PN sequence having a chip time of Tc. FIG. 8b represents a spread carrier and FIG. 8c represents a delayed spread carrier. FIG. 8d represents the sum of an original and a delayed signal assuming that the power of the time-delayed signal is comparable to the power of the original signal and that the delay is such that the two signals are in opposite phase at the receiver antenna. 20 FIG. 8e represents the spectrum of the resultant signal of FIG. 8d. If the power of the spread carrier and the delayed spread carrier is each A, then the power of the signal in FIG. 8d is 2A as shown at 810. In FIG. 8d the bulk of the signal has been canceled since the signals only add during chip transitions and then only for the duration of the time delay T_Del as shown in 810, 820, 830, 840 and 850. The resultant power of the signal shown in FIG. 8d is much smaller than the power of the original signal. If the power that would be available is P_TOT, then the power available at 830 or 840, for example, is ##EQU1## where T_Del is the delay time between the signals of FIGS. 8b and 8c. In a practical example for a chip rate of 5 million chips per second and a time delay of 20 nanoseconds, the power of the signal shown in FIG. 8d is 5 times smaller than the power of the original signal. In addition, the spectrum of the resultant signal is much wider, as shown in FIG. 8e and therefore only a part the total power of the signal can pass through the narrow band receiver filters. In the example discussed above, only one tenth of the energy originally present is available in the detector. Thus, the total loss of power, considering the above-described loss of 80% of the original power, is therefore approximately 50 times or 17 dB as shown at 880. The power at this lobe is P_TOT '/(T_c /T_Del).

The frequency shift arrangement described above provides for better suppression of the multipath effect because it introduces frequency diversity which changes the phase relationships of the original and time-delayed signals. This reduces or eliminates the cancellation effect shown in FIG. 8d. For a carrier frequency of approximately 1GHz, the total loss encountered may be less than 3 dB as opposed to 17 dB in the example above (FIG. 8e).

Another feature of the receiver is related to the reception of strong signals above the dynamic range of the receiver signal strength indicator (RSSI). In the case when the signal is very strong and considerably above the noise level, there may be no need to despread the signal before detection. According to FIG. 4, if the spectrum of each component of the spread spectrum signal is narrow enough (in practice, less than 1/Ts) the data can be recovered directly from any of the spectral components without prior despreading. This permits turning off the local PN generator when a predetermined signal strength level has been exceeded for a predetermined period of time thus disabling the acquisition process. Therefore the evaluation of the RSSI signal, which aids the acquisition, is irrelevant. The signal received is so strong that peaks of the spectral lines would be received without despreading. Each spectral lines carries the full modulation; thus allowing the data to be extracted from a single spectral line. This known technique has very practical implications since the RSSI amplifier usually saturates at signal levels which are much smaller than the maximum signal level at which the limiters and the FM discriminator can be expected to function properly. The effective dynamic range of the receiver can be extended in this manner without the necessity for automatic gain control circuitry. In addition, the acquisition time can be reduced for all transmitters which are close enough to the receiver to produce signals at the receiver antenna which are strong enough to actuate this detection mechanism. This type of a narrow band signal spectrum associated with each of the spectral lines can be obtained by using an appropriately chosen deviation together with a bit time Tb much greater than the PN sequence period Ts.

The signals generated in a preferred embodiment of the system according to the present invention are shown in FIG. 9. FIGS. 9A and 9B show the spectral envelope of the transmitted signal and its relationship to the Federal Communications Commission (FCC) modulation mask for spread spectrum transmission. The modulation mask allots the frequency band of 902 to 928 MHz and requires that the signal at the boundaries be approximately 20 dB below the level of the transmitted frequencies Ftx1 and FTx2. In FIG. 9, the following parameters were chosen:

M=128

P=2

L=32

Fr=7.12 MHz and

N=4

These choices result in component spacing of the spread signal of 111.25 kHz and a df of 111.25/(2*129) or approximately 430 Hz. The receiver reference oscillator and the transmitter reference oscillator frequencies are changed from 7.12 (T1) to 7.12043 MHz (T2) or by approximately 60 ppm. The spectra in FIGS. 9a and 9b follow the (sin x/x)² rolloff. The lobe 906 at the 902 MHz lower boundary 912 is approximately 19.5 db down from the lobe 902 at frequency Ftx1=911.36 MH_z during time T1. The lobe 924 at the 928 MHz upper boundary 926 is approximately 20.2 dB down from the lobe 920 at FxT2=918.44+129*df=918.496 MHz during time T2. Thus minimal filtering is required at the lower boundary and no filtering is required at the upper boundary to meet the FCC modulation mask requirements. This significantly reduces the complexity and cost of the transmitter. This permits the use of nonlinear Class C or similar amplifiers in the transmitter which results in reduced power consumption and thus makes the transmitter more suitable for battery operation.

FIGS. 9c and 9d show the close-in spectra within lines 930. As can be seen these spectra are separated by 111.25 KHz. Comparing FIG. 9d to 9c, the spectra of FIG. 9d are offset by approximately 1/2 chip=56 KHz. A receiver bandwidth (BW) of 50 kHz fulfills the requirement that BW is less than 1/(2Ts) and allows for the reception of the modulated signal at a data rate of 111.25/4 kilobits per second or approximately 27.8 kilobits per second.

While a particular embodiment of the present invention has been disclosed herewithin, certain changes and modifications will readily occur to those skilled in the art. For example, those skilled in the art will recognize that the frequencies chosen for the data transmission may differ from each other by odd multiples of L/2 * Ts provided one can accept some of the undesirable jamming signal being present in the receiver pass band. Thus dF may be an approximation of L/(2*Ts) where L is an odd integer, although the performance will be degraded. All such changes and modifications can be made without departing from the invention as defined by the appended claims.

Claims

1. A method of transmitting digital data, comprising the steps of:

a) generating a high frequency carrier at a first carrier frequency;

b) modulating the frequency of said carrier with data bits assembled in data packets of predetermined format;

c) generating a high frequency carrier at a first carrier frequency and shifting the nominal frequency of said carrier by a frequency increment df to obtain a second carrier frequency, where

DF=K*1/Ts+df,

dF=approximately L/(2*Ts),

K is an integer,

L is an odd integer, and

Ts is a pn (pseudo noise) sequence period;

d) modulating the frequency of said carrier in step c) with said data bits assembled in data packets of said predetermined format;

e) spreading said carrier at said first and said second frequency, during a transmission time, by a predetermined pn sequence having a period Ts related to a bit time Tb where

Tb=N * Ts, N being an integer >1 whereby a spread carrier spectrum comprising many spectral components separated by 1/Ts is obtained, the amplitudes of said components being reduced by said spreading function, said components and the modulation imposed upon them being separable from other components by a narrow band filter.

2. The method according to claim 1, further comprising the steps of:

f) receiving the transmitted data; and

g) multiplying said transmitted spread carrier in a receiver by a locally generated predetermined pn sequence, for collapsing the bandwidth of the received spread carrier when the local pn sequence phase is in agreement with the sequence phase imposed on the received spread carrier thereby spreading any jamming signals which are received along with the desired transmitted signal into many components separated by 1/Ts intervals and reduced in amplitude by the spreading function.

3. The method according to claim 1, wherein step e) generates spectral components of the spread carrier that are separated by a frequency interval greater than 2 * IFbw, where Ifbw is the receiver bandwidth after step g).

4. The method according to claim 1, wherein a chip clock producing a signal at a frequency of 1/Ts 1/Tc and the carrier frequency are synchronously derived from one frequency source.

5. The method according to claim 4, wherein the data frequency is synchronously derived from said one frequency source.

6. The method according to claim 1, wherein K is greater than a sequence length for generating a frequency shift substantially equal to or greater than a chip sequence frequency of 1/Ts.

7. The method according to claim 1, wherein said predetermined data packet format comprises a preamble period during which said carrier frequency is equal to a nominal frequency and a data period during which the carrier frequency is modulated by said data bits.

8. The method according to claim 1 wherein each of said components are modulated in the same manner as said carrier prior to spreading and whereby the resulting modulated component spectra do not overlap.

9. A transmitter for transmitting digital data comprising:

a) carrier frequency generator means for generating a carrier frequency signal at a first frequency.;

b) modulator means for modulating said carrier with data bits;

c) frequency shift means for generating a second carrier frequency offset from said first carrier frequency by df, where

DF=K*1/Ts+df,

dF=approximately L/(2*Ts),

K is an integer,

L is an odd integer, and

Ts is a pn (pseudo noise) sequence period;

d) pseudo random sequence generator means for generating a predetermined pn sequence;

d) carrier spread means for spreading power of said first and second earlier frequencies by said pseudo noise sequence.

10. The transmitter according to claim 9 further including a phase lock loop means for deriving said first carrier frequency from a crystal reference oscillator.

11. The transmitter according to claim 10 further including divider means in said phase lock loop for dividing said first carrier frequency by M or M+1, where M is an integer; means for generating a frequency control signal, said frequency control signal controlling said divider means to divide by M or M+1.

12. The transmitter according to claim 11 wherein M is equal to 64.

13. The transmitter according to claim 11 wherein M is equal to 128.

14. The transmitter according to claim 9 wherein said frequency shift means generates said second carrier at a frequency shift from said first carrier frequency equal to or greater than a chip sequence frequency of 1/Ts.

15. The transmitter according to claim 9 wherein said transmitter further incudes means for deriving chip frequency of 1/Ts 1/Tc and said first carrier frequency from a single reference frequency source.

16. The apparatus according to claim 9 wherein said transmitter further includes means for deriving chip frequency of 1/Ts 1/Tc, said first carrier frequency and a data clock frequency from a single reference frequency source.

17. The transmitter according to claim 9 wherein said transmitter further includes means for generating said formatted data packet for modulating said carrier frequency, said data packet comprising a preamble period during which the carrier frequency is equal to a nominal value and data period during which the carrier frequency is deviated according to the corresponding data bit values.

18. A receiver for a spread spectrum signal comprising:

a) a wideband receiving means for receiving a spread carrier radio signal for generating a spread carrier electrical signal;

b) means for generating a predetermined pn (pseudo noise) sequence;

c) multiplying means coupled to said receiving means and said generating means for multiplying said spread carrier electrical signal by said predetermined pn sequence, for collapsing a bandwidth of the received spread carrier when the local pn sequence phase is in agreement with the sequence phase imposed on the received spread carrier by a transmitter means, thereby spreading any jamming signals which are received along with the transmitted spread signal into many components separated by 1/Ts intervals and reduced in amplitude by the spreading function;

d) narrow band FM receiver means coupled to an output of said multiplying means for recovering data;

e) means coupled to said wideband receiving means for alternately receiving a radio signal on one of two preselected frequencies; and

f) frequency switch means coupled to said means for alternately receiving for switching the received frequency between a first and a second of said preselected frequencies at predetermined time intervals.

19. The receiver according to claim 18 wherein a bandwidth said narrow band FM receiver means is less than 1/(2*Ts) where Ts is a period of said pn sequence.

20. The receiver according to claim 19 wherein said means for alternately receiving is a local oscillator and said frequency switch means changes the frequency thereof.

21. The receiver according to claim 19 wherein said means for alternately receiving comprises an independent means for receiving operating at a second one of said preselected frequencies.

22. The receiver according to claim 18 wherein each of said receiver means further includes means for deriving a chip frequency of 1/Ts 1/Tc and a carrier frequency from a single reference frequency source.

23. The receiver according to claim 22 wherein each of said receiver means further includes means for deriving a chip frequency of 1/Ts 1/Tc, a carrier frequency, and a data clock frequency from a single reference frequency source.

24. The receiver according to claim 22 wherein each of said receiver means further includes means for automatic frequency control (AFC) and means for activating said AFC upon synchronization of a phase of said pn sequence with a phase of the pn sequence modulating said received signal.

25. The receiver according to claim 18 wherein each of said receiver means further includes signal strength indicator means to detect power of said received signal.

26. A system for transmitting and receiving digital data, comprising:

a) means for generating a high frequency carrier at a first carrier frequency;

b) means for modulating the frequency of said carrier with data bits assembled in data packets of predetermined format;

c) means for shifting the nominal frequency of said carrier by a frequency increment df to obtain a second carrier frequency, where

DF=K*1/Ts+df,

dF=approximately L/(2*Ts),

K is an integer,

L is an odd integer, and

Ts is a pn (pseudo noise) sequence period;

d) means for spreading said carrier at said first and said second frequency, during a transmission time, by a predetermined pn sequence having a period Ts related to a bit time Tb where

Tb=N * Ts, N being an integer >1 whereby a spread carrier spectrum comprising many spectral components separated by 1/Ts is obtained, the amplitudes of said components being reduced by said spreading function,

e) means for receiving the transmitted data; and

f) means for multiplying said transmitted spread carrier in said receiver means by a locally generated predetermined pn sequence, for collapsing the bandwidth of the received spread carrier when the local pn sequence phase in agreement with the sequence phase imposed on the received spread carrier.

27. The system according to claims 26 further comprising:

g) narrow band FM receiver means coupled to an output of said multiplying means for recovering data;

h) means coupled to said receiving means for alternately receiving a radio signal on one of two preselected frequencies; and

i) frequency switch means coupled to said means for alternately receiving for switching the received frequency between a first and a second of said preselected frequencies at predetermined time intervals.

28. The system according to claim 27, wherein said means for spreading said carrier generates spectral components of the spread carrier that are separated by a frequency interval greater than 2 * IFbw, where IFbw is the receiver bandwidth after said multiplying means.

29. The system according to claim 26, wherein a chip clock producing a signal at a frequency of 1/Ts1/Tc and the carrier frequency are synchronously derived from the frequency source.

30. The system according to claim 29, wherein the data frequency is synchronously derived from said one frequency source.

31. The system according to claim 26, wherein K is greater than a sequence length for generating a frequency shift substantially equal to or greater than a chip sequence frequency of 1/Ts.

32. The system according to claim 26 wherein said predetermined data packet format comprises a preamble period during which said carrier frequency is equal to a nominal frequency and a data period during which the carrier frequency is modulated by said data bits.

33. The system according to claim 26 further comprising means for repeating the transmission of said data.

34. The system according to claim 26 wherein each of said components are modulated in the same manner as said carrier prior to spreading and whereby the resulting modulated component spectra do not overlap.

35. In a spread spectrum transmitter having a phase-lock loop for generating a carrier signal at a first frequency, said phase-lock loop including a voltage controlled oscillator having an output coupled to a frequency divider and an input coupled to an output of a phase detector, said phase detector having one input coupled to the output of said divider and a second input coupled to an output of a reference oscillator; frequency shift means for generating a second carrier frequency offset from said first carrier frequency; and means for generating a spread spectrum signal from said first and said second carrier and a spreading signal, the improvement comprising said frequency driver divider consisting of a dual modulus prescaler.

36. In a spread spectrum transmitter receiver having a phase-lock loop for generating a carrier signal at a first frequency, said phase-lock loop including a voltage controlled oscillator having an output coupled to a frequency divider and an input coupled to an output of a phase detector, said phase detector having one input coupled to the output of said divider and a second input coupled to an output of a reference oscillator; means for receiving a second carrier frequency offset from said first carrier frequency; means for generating a spread spectrum signal from said first and said second carrier and a spreading signal; and means for despreading an incoming rf signal utilizing said spread spectrum signal, the improvement comprising said frequency divider consisting of a dual modulus prescale prescaler.