Semiconductor integrated circuit with voltage limiter having different output ranges from normal operation and performing of aging tests

Abstract

In a voltage converter which is disposed in a semiconductor integrated circuit so as to lower an external supply voltage and to feed the lowered voltage to a partial circuit of the integrated circuit, the voltage converter is constructed so as to produce an output voltage suited to an ordinary operation in the ordinary operation state of the semiconductor integrated circuit and an aging voltage in the aging test of the circuit.

Description

This application is a continuation of application Ser. No. 562,969, filed December 19, 1983, now abandoned.

This is a reissue of application Ser. No. 140,628, filed Jan. 4, 1988, now U.S. Pat. No. 4,916,389, which is a Continuation of application Ser. No. 562,969, filed Dec. 19, 1983, now abandoned, which is a Continuation-In-Part of application Ser. No. 368,162, filed Apr. 14, 1982, now U.S. Pat. No. 4,482,958.

BACKGROUND OF THE INVENTION

The present invention relates to a voltage converter which lowers an external supply voltage within a semiconductor integrated circuit chip to drive circuits on the chip having small geometries.

Reduction in the geometries of devices such as bipolar or MOS transistors has been accompanied by a lowering in the breakdown voltages of the devices, which has made it necessary to lower the operating voltage of small geometry devices with an integrated circuit. From the viewpoint of users, however, a single voltage source of for example 5 V which is easy to use is desirable. As an expedient for meeting such different requests of IC manufacturers and the users, it is considered to be necessary to lower the external supply voltage V_CC within a chip and to operate the small geometry devices with the lowered voltage V_L.

FIG. 1 shows an example of such an expedient, in which the circuit A' of the whole chip 10 including, e.g., an input/output interface circuit is operated with the internal supply voltage V_L lowered by a voltage converter 13.

FIG. 2 shows an integrated circuit disclosed in Japanese Patent Application No. 56-57143 (Japanese Laid-open Patent Application No. 57-172761). The small geometry devices are employed for a circuit A determining the substantial density of integration of the chip 10, and are operated with the voltage V_L obtained by lowering the external supply voltage V_CC by means of a voltage converter 13. On the other hand, devices of comparatively large geometries are employed for a driver circuit B including, e.g., an input/output interface which does not greatly contribute to the density of integration which are operated by applying V_CC thereto. Thus, a large-scale integrated circuit (hereinbelow, termed "LSI") which operates with V_CC when viewed from outside the chip becomes possible.

However, when such an integrated circuit is furnished with the voltage converter, an inconvenience is involved in an aging test which is performed after the final fabrication step of the integrated circuit.

The terminology "aging test" as used herein identifies a test performed after the final fabrication step of the integrated circuit during which voltages higher than in an ordinary operation are intentionally applied to the respective transistors in the circuit to test the integrated circuit for break down due to an inferior gate oxide film.

The aforementioned voltage converter in Japanese Patent Application No. 56-57143 functions to feed the predetermined voltage. Therefore, the circuit fed with the supply voltage by the voltage converter cannot be subjected to the aging test.

In order to solve this problem, an invention disclosed in U.S. Pat. No. 4,482,985 has previously been made, but it has had difficulty in the performance for actual integrated circuits. As illustrated in FIGS. 2 to 6 in that patent, according to the cited invention, an internal voltage increases up to an aging point rectilinearly or with one step of change as an external supply voltage increases. Accordingly, the internal voltage changes greatly with the change of the external supply voltage. This has led to the disadvantage that the breakdown voltage margins of small geometry devices in an ordinary operation become small.

SUMMARY OF THE INVENTION

An object of the present invention is to further advance the invention disclosed in U.S. Pat. No. 4,482,985 referred to above, and to provide a voltage converter which can widen the margins of the breakdown voltages of small geometry devices in an ordinary operation and which affords sufficient voltages in an aging test.

The present invention consists in that the output voltage of a voltage converter is set at a voltage suitable for the operations of small geometry devices against the change of an external supply voltage when a semiconductor integrated circuit is in its ordinary operation region, and at an aging voltage when the ordinary operation region is exceeded.

To this end, according to the voltage converter of the present invention, when the external supply voltage has been changed from the lower limit value of the ordinary operation range thereof to the aging operation point thereof, the output voltage of the voltage converter changes up to the aging voltage without exhibiting a constant changing rate versus the change of external supply voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1 and 2 show semiconductor integrated circuits each having a voltage converter.

FIGS. 3 and 5 show basic circuits each of which constitutes a device embodying the present invention.

FIGS. 4 and 6 show the characteristics of the circuits in FIGS. 3 and 5, respectively.

FIGS. 7, 9 and 11 show devices embodying the present invention.

FIGS. 8, 10 and 12 show the characteristics of the circuits in FIGS. 7, 9 and 11, respectively.

FIGS. 13(A) to (C) and 14(A) to (C) show prior art voltage regulators and FIGS. 13B and 14B show the formation of the circuit of FIG. 3 in practicable forms and characteristics of such practicable forms.

FIG. 15 shows the characteristic in FIG. 4 more specifically.

FIG. 16 shows another practicable example of the circuit in FIG. 3.

FIG. 17 shows the characteristic in FIG. 8 concretely.

FIG. 18 shows a circuit for producing the characteristics in FIG. 17.

FIG. 19 shows the characteristic in FIG. 8 concretely.

FIG. 20 shows a circuit for producing the characteristics in FIG. 19.

FIG. 21 shows the characteristic in FIG. 10 concretely.

FIG. 22 shows a circuit for producing the characteristic in FIG. 21.

FIG. 23 shows a characteristic in another embodiment of the present invention.

FIG. 24 shows a circuit for producing the characteristic in FIG. 23.

FIG. 25 shows the characteristic in FIG. 12 concretely.

FIG. 26 shows a circuit for producing the characteristic in FIG. 25.

FIG. 27 shows a practicable example of the circuit in FIG. 26.

FIG. 28 shows the actual characteristics of the circuit in FIG. 27.

FIG. 29(A) shows a gate signal generator for use in an embodiment of the present invention.

FIG. 29(B) shows a time chart of the circuit in FIG. 29(A).

FIG. 30 shows a protection circuit which connects the circuit of FIG. 29(A) with the circuit of FIG. 16, 18, 20, 22, 24, or 26.

FIG. 31 shows a practicable circuit of an inverter for use in the circuit of FIG. 29(A).

FIG. 32 shows a practicable circuit of an oscillator for use in the circuit of FIG. 29(A).

FIG. 33 shows an example of a buffer circuit for the output of the circuit shown in FIG. 16, 18, 20, 22, 24 or 26.

FIG. 34 shows the characteristics of the circuit in FIG. 33.

FIGS. 35, 36 and 37 show other examples of buffer circuits, respectively.

FIG. 38 shows a time chart of the circuit in FIG. 37.

FIG. 39 shows a practicable example of the circuit in FIG. 3.

FIG. 40 shows an example of a buffer circuit.

FIG. 41 shows the characteristics of the circuit in FIG. 40.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Voltage converter circuit forms for affording various output characteristics versus an external supply voltage V_CC, as well as practicable examples thereof, will be first described, followed by practicable embodiments on a method of feeding power to the voltage converter and on a buffer circuit for the voltage converter well suited to drive a large load.

FIGS. 3 and 5 show basic circuits which are used for forming voltage converter embodiments of the present invention for providing a voltage V_L to circuits such as shown in FIGS. 1 and 2.

In the circuit of FIG. 3, a resistance R₃ in FIG. 23 of U.S. Pat. No. 4,482,985 is replaced by a variable impedance arrangement described below, and a transistor Q is employed in order to enhance a current driving ability for a load to which an output voltage V_L is applied. Here, the control terminal voltage V_G of the transistor Q has a characteristic which changes versus the change of an external supply voltage V_CC and which is the output voltage of a reference voltage generator REF. More specifically, as illustrated in FIG. 4, in a case where the external supply voltage V_CC is gradually increased from 0 (zero) V, the voltage V_G rises abruptly when a certain voltage V_P has been reached, so that the transistor Q turns "on". For V_CC not smaller than V_P, Q continues to turn "on". Therefore, the effective impedance of the whole basic circuit BL decreases, and the ratio thereof with the effective impedance R changes, so that the voltage V_L becomes a straight line of different slope for V_CC not smaller than V_P as shown in FIG. 4. Here in FIG. 4, the example is illustrated in which V_G rises abruptly from O V to a certain voltage for V_CC not smaller than V_P. However, it is also allowed to adopt a characteristic in which, in case of changing V_CC from O V, V_G rises gradually from O V and becomes, at the point V_P, a voltage level to turn "on" the transistor Q. Regarding the example in which V_G rises abruptly at and above the certain voltage V_CC, the reference voltage generator can be realized by the cascade connection of devices having rectification characteristics as taught in U.S. Pat. No. 4,482,985. Regarding the example in which V_G rises gradually, the reference voltage generator can be realized by a simple resistance divider circuit. In FIG. 4, the coefficient of V_L relative to V_CC can be changed at will by the designs of the resistance and the transistor Q.

FIG. 5 shows another example which employs the same basic circuit BL as in FIG. 3. Whereas the example of FIG. 3 derives V_L from the V_CC side, this example derives V_L from the ground side. When the characteristic of the output voltage V_G from the reference voltage generator is set in advance so that the transistor Q may turn "on" at V_CC not smaller than V_P, V_L is determined by the effective impedance of the whole basic circuit BL and the effective impedance R, and hence, V_L becomes as shown in FIG. 6.

While FIGS. 3 and 5 have exemplified the transistors as being MOS transistors, bipolar transistors may be used if desired. Particularly in a case where whole chips are constructed of MOS transistors in the examples of FIGS. 1 and 2, it is usually easier to design them when the circuits of FIGS. 3 and 5 are constructed of MOS transistors. In a case where the whole chips are of bipolar transistors, it is more favorable to use bipolar transistors. It is sometimes the case, however, that the chip includes both MOS transistors and bipolar transistors. It is to be understood that, in this case, the MOS transistor or/and the bipolar transistor can be used for the circuit of FIG. 3 or FIG. 5 in accordance with an intended application. In addition, although the examples of FIGS. 4 and 6 have been mentioned as the characteristics of the circuit REF, these examples are not especially restrictive, but the characteristic of the circuit REF may be set according to the purpose of the design of V_L.

Now, a voltage converter based on the circuit of FIG. 3 will be described. FIGS. 7 and 8 illustrate an example in which the basic circuits BL numbering k are connected in parallel with the effective impedance R of the circuit of FIG. 3 (formed by the resistor and the basic circuit BL₀). Each of the basic circuits BL corresponds in structure to the basic circuit BL shown in FIG. 3, but are respectively set to turn on their transistors Q at different levels of the supply voltage V_CC. For example, the circuits REF in the respective basic circuits BL are set so that BL₀ may first turn "on" at V_P0, BL₁ may subsequently turn "on" at V_P1, and BL_k may lastly turn "on" at V_Pk as shown in FIG. 8. The transistors in the respective circuits BL are designed so that the coefficients of the changes of the respective voltages V_L versus the voltage V_CC may be varied. As V_CC increases more, impedances are successively added in parallel with the impedance R of the resistor and the basic circuit BL₀, so that the entire characteristic of V_L becomes concave for V_CC not smaller than V_P0.

The coefficients of the changes are varied for the following reason. For example, in a case where the aging operation points are V_P2, V_P3, . . . and V_Pk and where the aging voltages of circuits to be fed with the supply voltages by the voltage converter are V_L2, V_L3, . . . and V_Lk, the transistion is smoothed when the first aging operation point shifts to the next one.

The present circuit is a circuit which is practical in terms of the operating stability of the ordinary operation and an effective aging for the system of FIG. 2. By way of example, the V_CC operation point in the ordinary operation is set at a point at which V_L changes versus V_CC as slightly as possible, that is, the coefficient of change is the smallest, in order to achieve a stable operation. In fact, if desired, the coefficient of change of V_L versus V_CC can be set to be zero in the range between V_P1 and V_P2 for the ordinary operation so that a constant voltage V_L is held in this entire range. Alternatively, a small positive slope can be used in this, as shown in FIG. 8. On the other hand, the V_CC operation point in the aging test is set at a point at which the coefficient of change is great, in order to approximately equalize the stress voltage conditions of transistors of large geometries receiving V_CC to stress voltage conditions of transistor of small geometries receiving V_L as described in, U.S. Pat. No. 4,482,985. Specifically, large geometry devices such as those found in the interface circuit B of FIG. 2 are operated during aging tests at a higher potential than small geometry devices in circuit A at the reduced potential produced by voltage converter 13. More concretely, in case of using only BL₀ and BL₁ in the circuit of FIG. 7, the coefficient of change in FIG. 8 may be made small between the lower limit voltage V_P0 (e.g., 2-3 V) and the upper limit voltage V_P1 (e.g., 6 V), to set the ordinary operation point (e.g., 5 V) concerning V_CC for the ordinary operation range in this section, while the coefficient of change may be made great between V_P1 and V_P2 (e.g., V_P2 being 7-9 V), to set the aging operation point (e.g., V_CC =8 V) in this section. The ordinary operation range is solely determined by ratings, and it is usually set at 5±0.5 V. It is to be understood that, for some purposes of designs, the operation voltage points and the aging voltage points can be set at any desired V_CC points by employing the basic circuits BL₂, BL₃ . . . etc. When more circuits BL are used, the V_L characteristic can also be made smoother versus V_CC, so that the operation of the internal circuit can be stabilized more. Further, since the V_CC voltage is high in the aging test, it is effective to construct the voltage converter itself using high breakdown voltage transistors. To this end, the voltage converter may be constructed of transistors of large geometry in the system of FIG. 2 by way of example.

FIGS. 9 and 10 show an example of using the FIG. 4 arrangement with additional basic circuits BL being connected in parallel on the ground side. In this arrangement, by setting the respective circuits BL to have different turn-on times, the characteristic of the whole V_L can be made convex relative to V_CC, as shown in FIG. 10. This characteristic is effective for protecting the circuit A' from any overvoltage V_L in the system of FIG. 1 by way of example. This achieves the advantage that, in case of measuring the V_CC voltage margin of the whole chip, a sufficiently high voltage V_CC can be applied without destroying small geometry devices.

In some uses, it is also possible that the circuits of FIGS. 7 and 9 can coexist. By way of example, the ordinary operation point is set at a point at which the coefficient of change is small, and the aging operation point is set at a point at which the coefficient of change is great. These are realized by BL₀ and BL₁ in the circuit of FIG. 7. Further, in order to make the coefficient of change small again at and above the V_CC point of the aging condition to the end of preventing the permanent breakdown of devices, the basic circuits BL other than BL₀ are connected so as to operate in parallel with the latter as in the circuit form of FIG. 9. This makes it possible to design a circuit in which the devices are difficult to break down at and above the V_CC point of the aging operation.

Thus, even when the supply voltage has erroneously been made abnormally high, by way of example, the breakdown of the devices can be prevented.

FIGS. 11 and 12 show an example in which a basic circuit BL' is connected in parallel with the circuit of FIG. 3, whereby the changing rate of V_L is made negative at and above V_P ' which is a certain value of V_CC. More specifically, when V_CC is increased, the transistor Q first turns "on" while the output voltage V_G of the reference voltage generator 1 in the basic circuit BL is not lower than V_P, so that the gradient of V_L versus V_CC decreases. A reference voltage generator 2 is designed so that a transistor Q' in the basic circuit BL' may subsequently turn "on" at the certain V_CC value, namely, V_P '. In addition, the conductance of Q' is designed to be sufficiently higher than that of Q. Then, the V_L characteristic after the conduction of the transistor Q' is governed by the characteristic of BL', so that V_L comes to have the negative gradient as shown in FIG. 12.

The merit of the present circuit is that, when the aforementioned point at which V_L lowers is set at or below the breakdown voltages of small geometry devices, these small geometry devices are perfectly protected from breakdown even when the voltage V_CC has been sufficiently raised. For example, a measure in which the output voltage V_L lowers when a voltage higher than the external supply voltage V_CC at the aging point has been applied is especially effective because any voltage exceeding the aging point is not applied to the devices.

It is to be understood that an external instantaneous voltage fluctuation can also be coped with.

Obviously, the circuit of FIG. 5 can afford any desired V_L characteristic by connecting the basic circuit BL' in parallel as in the example of FIG. 3.

While, in the above, the conceptual examples of the voltage converters have been described, practicable circuit examples based on these concepts will be stated below.

FIG. 13A shows an example of the circuit of FIG. 3 which employs a bipolar transistor. A voltage regulator circuit CVR is, for example, a cascade connection of Zener diodes or ordinary diodes the terminal voltage of which becomes substantially constant. FIG. 13(A) indicates a well-known voltage regulator which has the characteristic shown by (A) in FIG. 13(C). This voltage regulator is described in detail in "Denpa-Kagaku (Science of Electric Wave)", February 1982, p. 111 or "Transistor Circuit Analysis", by Joyce and Clarke, Addison-Wesley Publishing Company, Inc., p. 207. Since, however, V_L is a fixed voltage in this condition, a resistance r can be connected in series with the CVR as shown in FIG. 13(B) in accordance with the present invention to slope the curve as desired. Thus, V_L comes to have a slope relative to V_CC as shown by the characteristic (B) shown in FIG. 13(C).

FIG. 14 shows another embodiment. FIG. 14(A) indicates a well-known voltage regulator which employs an emitter follower and which has the characteristic shown by (A) in FIG. 14(C). Since V_L is also a fixed voltage, a resistance r is used in FIG. 14(B) in order to provide a desired slope. Thus, a characteristic as shown as characteristic (B) in FIG. 14(C) is provided.

These examples of FIGS. 13 and 14 are especially suited to the system as shown in FIG. 1. In FIG. 1, usually a great current flows through the circuit associated with the input/output interface. Therefore, a high current driving ability is required of the voltage converter correspondingly. Obviously, the voltage converter constructed of the bipolar transistor is suited to this end.

Next, there will be explained practicable examples in which voltage converters are constructed of MOS transistors on the basis of the circuits of FIGS. 3, 7, 9 and 11.

FIG. 15 shows a concrete example of the characteristic of FIG. 4 in which V_L is endowed with a slope m for V_CC of and above a certain specified voltage V₀. Since the change of V_L decreases for the voltage not smaller than V₀, the breakdown of small geometry devices is less likely to occur to that extent.

V_L =V_CC is held for V_CC smaller than V₀, for the following reason. In general, MOSTs have their operating speeds degraded by lowering in the threshold voltages thereof as the operating voltages lower. To the end of preventing this drawback, it is desirable to set the highest possible voltage on a lower voltage side such as V_CC smaller than V₀. That is, V_L should desirably be equal to V_CC.

FIG. 16 shows an embodiment of a practicable circuit DCV therefor, which corresponds to a practicable example of the circuit of FIG. 3.

The features of the present circuit are that the output voltage V_L is determined by the ratio of the conductances of MOS transistors Q₀ and Q_l, and that the conductance of the MOS transistor Q_l is controlled by V_L.

With the present circuit, letting the gate voltage V_G of Q_O be V_CC +V_th(O) (where V_th(O) denotes the threshold voltage of the MOST Q_O), the control starting voltage V_O and the slope m are expressed as follows: ##EQU1## Here, β(O) and β(1) denote the channel conductances of Q_O and Q_l, V_th(i) (i=1-n) and V_th(l) denote the threshold voltages of the MOS transistors Q_i (i=1-n) and Q_l, and n denotes the number of stages of Q_i.

Accordingly, V_O and m can be varied at will by n, V_th(i), V_th(l) and β(1)/β(O). It has been stated before that V_L =V_CC is desirable for V_CC smaller than V_O. In this regard, for V_CC smaller than V_O, V_L is determined by V_O because Q_l is "off". Therefore, the voltage V_G of Q_O must be a high voltage of at least V_CC +V_th(O).

In order to simplify the computation and to facilitate the description, the circuit of FIG. 16 is somewhat varied from an actual circuit. As a practical circuit, as shown in FIG. 27 to be referred to later, a transistor of similar connection (Q_S(1.6) in FIG. 27) needs to be further connected between the n-th one of the transistors connected in cascade and the ground. That is, a kind of diode connection is made toward the ground. With this measure, when V_CC has been varied from the high voltage side to the low voltage side, the nodes of the transistors connected in cascade are prevented from floating states to leave charges behind. For the sake of the convenience of the description, the transistor of this measure shall be omitted in the ensuing embodiments.

FIG. 17 shows a characteristic in which, when the external supply voltage V_CC changes between the lower limit value V_O and upper limit value V_O ' of the ordinary operation range, the slope m of the output voltage V_L is small, and a slope m' which corresponds to the external supply voltage greater than V_O ' is made steeper than m.

FIG. 18 shows an example of a circuit for producing the characteristic of FIG. 17.

These correspond to a practicable form of the example of FIGS. 7 and 8.

The feature of the present circuit is that, between the terminals 1 and 2 of the circuit DCV shown in FIG. 16, a circuit DCV2 similar to DCV1 is added, whereby the conductance of a load for DCV1 is increased at and above V_O ' so as to increase the slope of V_L.

With the present circuit, the second control starting voltage V_O ' is expressed by: ##EQU2## In addition, the slope m' is determined by the ratio between the sum of the conductances of the MOS transistors Q_O and Q'_l and the conductance of the MOS transistor Q_l. Here, V'_th(i) (i=1-n') and V'_th(l) denote the threshold voltages of the MOS transistors Q'_i (i=1-n') and Q'_l, respectively.

Accordingly, V'_O and m' can be varied at will by n, n', β(1), β'(1), V_th(i), V_th(l), V'_th(i) and V'_th(l). Here, β'(1) denotes the channel conductance of the MOS transistor Q'_l.

This circuit has the ordinary operation range between the lower limit value V_O and the upper limit value V_O ', and is effective when the aging point has a value larger than V_O '. That is, since the slope m is small in the ordinary operation region, margins for the breakdown voltages of small geometry devices are wide, and power consumption does not increase. Here, the slope m' for the external supply voltage higher than the ordinary operation region is set for establishing a characteristic which passes an aging voltage (set value).

In an example illustrated in FIG. 19, a characteristic in which the slope of V_L becomes m" greater than m' when the external supply voltage V_CC has reached V_O " is further added to the characteristic shown in FIG. 17.

FIG. 20 shows an example of a practicable circuit therefor. These correspond to a concrete form of the example of FIGS. 7 and 8. The feature of the present circuit is that circuits DCV2 and DCV3 similar to the circuit DCV1 are added between the terminals 1 and 2 of the circuit shown in FIG. 16, whereby the conductance of the load for DCV1 is successively increased so as to increase the slope of V_L in two stages at the two points V_O ' and V_O ".

With the present circuit, the second and third control starting voltages V_O ' and V_O " are respectively expressed by: ##EQU3## Here, V"_th(i) (i=1-n") and V"_th(l) denote the threshold voltages of the MOS transistors Q"_i (i=1-n") and Q"_l, respectively. Besides, the slope m' is determined by the ratio between the sum of the conductances of the MOS transistors Q_O and Q'_l and the conductance of the MOS transistor Q_l, and the slope m" by the ratio between the sum of the conductances of the MOS transistors Q_O, Q'_l and Q"_l and the conductance of the MOS transistor Q_l.

Accordingly, V_O ' and m' can be varied at will by n, n', β(O), β(l), β'(l), V_th(i), V_th(l), V'_th(i) and V'_th(l), while V"_O and m" by n, n', n", β(O), β(l), β'(l), β"(l), V_th(i), V_th(l), V'_th(i), V'_th(l), V"_th(i) and V"_th(l). Here, β"(l) denotes the channel conductance of Q"_l.

This circuit is effective when the ordinary operation range extends between the lower limit value V_O and the upper limit value V_O ', and aging tests are carried out in the two sections of the external supply voltage V_CC ≧V_O " and V_O '<V_CC <V_O ". The aging tests in the two sections consist of the two operations: aging for a short time, and aging for a long time. The former serves to detect a defect occurring, for example, when an instantaneous high stress has been externally applied, while the latter serves to detect a defect ascribable to a long-time stress.

FIG. 21 shows an example wherein, when the external supply voltage V_CC is greater than V_O ', the slope m' of the voltage V_L is set at m>m' under which the output voltage V_L follows up the external supply voltage V_CC.

FIG. 22 shows an embodiment of a practicable circuit therefor. These correspond to a concrete form of the example of FIGS. 9 and 10. The feature of the present circuit is that a circuit DCV2 similar to DCV1 is added between the terminal 2 and ground of the circuit shown in FIG. 16, whereby the conductance of a load for the transistor Q_O is increased at V_O ' so as to decrease the slope of V_L.

With the present circuit, the second control starting voltage V_O ' is expressed by: ##EQU4## In addition, the slope m' is expressed by the ratio between the conductance of Q_O and the sum of the conductances of Q_l and Q'_l.

Accordingly, V_O ' and m' can be varied at will by n, n', β(O), β(l), β'(l), V_th(i), V_th(l), V'_th(i) and V'_th(l).

This circuit is applicable to devices of lower breakdown voltages. Usually, when the breakdown voltages of devices are low, the output voltage V_L of the ordinary operation region (V_O <V_CC <V_O ') may be suppressed to a low magnitude. In some cases, however, the magnitude of V_L cannot be lowered because the operating speeds of a circuit employing small geometry devices and a circuit employing large geometry devices are matched. In such cases, the slope m_a of the output voltage V_L is decreased in order for the aging operation point to be passed. Thus, the magnitude of the output voltage V_L can be raised near to the withstand voltage limit of the devices within the range of the ordinary operation region, and the operating speed of the circuit employing the small geometry devices can be matched with that of the circuit employing the large geometry devices.

In an example shown in FIG. 23, a characteristic in which the slope of V_L becomes m" smaller than m' when the external supply voltage C_CC has reached V_O " is further added to the characteristic illustrated in FIG. 17.

FIG. 24 shows an embodiment of a practicable circuit therefor. This corresponds to an example in which the examples of FIGS. 7 and 9 coexist. The feature of the present circuit is that the embodiments of FIGS. 18 and 21 are combined thereby to increase and decrease the slope of V_L at the two points V_O ' and V_O " respectively.

With the present circuit, the second and third control starting voltages V_O ' and V_O " are respectively expressed by: ##EQU5## In addition, the slope m' is expressed by the ratio between the sum of the conductances of Q_O and Q_l ' and the conductance of Q_l, while m" is expressed by the ratio between the sum of the conductances of Q_O and Q_l ' and the sum of the conductances of Q_l and Q_l ".

Accordingly, V_O ' and m' can be varied at will by n, n', β(O), β(l), β'(l), V_th(i), V'_th(i) and V'_th(l), while V_O " and m" can be varied by n, n', n", β(O), β(l), β'(l), β"(l), V_th(i), V_th(l), V'_th(i), V'_th(l), V"_th(i) and V"_th(l).

This circuit protects small geometry devices from permanent breakdown in such a way that, even when V_CC has become higher than the withstand voltage limit V_O " of the devices due to some fault of the external power source, it does not exceed a breakdown voltage V_B. That is, the slope m" of V_L for V_CC not smaller than V_O " is made gentler than the slope m' in the aging, whereby even when the external supply voltage V_CC has become V_O " or above, the output voltage V_L is prevented from exceeding the breakdown voltage (usually, higher than the withstand voltage limit) of the devices. This makes it possible to prevent the device breakdown even when the supply voltage has been raised abnormally by way of example.

FIG. 25 shows an example in which the slope m' is made negative when the external supply voltage V_CC has exceeded V_O '.

FIG. 26 shows an embodiment of a practical circuit therefor. These correspond to a concrete form of the example of FIGS. 11 and 12. The feature of the present circuit is that the drain of Q₁ ' in DCV2 is connected to the terminal 1 of the circuit shown in FIG. 16, the drain of Q_l ' to the terminal 2, and the source of Q_l ' to the ground, whereby the conductance of Q_l ' is controlled by V_CC, and besides, it is made greater than the conductance of Q_O so as to establish m'<O.

With the present circuit, the second control starting voltage V_O ' and the slope m' are expressed by the following on the assumption of β'(l)≧β(O): ##EQU6##

Accordingly, V_O ' and m' can be varied at will by n', V'_th(i), V'_th(l) and β'(l)/(O).

FIGS. 27 and 28 show a practicable example of the present circuit and examples of the characteristics thereof. All the threshold voltages of transistors are 1 (one) V, and V_G =V_CC +V_th(O) is held. In addition, numerals in parentheses indicate values obtained by dividing the channel widths by the channel lengths of the transistors. FIG. 28 illustrates V_L with a parameter being the corresponding value W_l /L_l of Q_l '. By way of example, the voltage in the ordinary operation is set at 5 V, and the aging voltage at 8 V.

This circuit consists in that the slope of the voltage at and above V_O " in the characteristic shown in FIG. 23 is made negative, thereby to intensify the aspect of the device protection of the circuit in FIG. 24.

With this circuit, the breakdown due to the external application of a high voltage is perfectly prevented, and the power consumption in the integrated circuit does not exceed an allowable value. Thus, even when the instantaneous high voltage has been externally applied, the prevention of the breakdown of the devices is ensured.

Thus far, the voltage converters and their characteristics have been described. Next, the method of feeding the voltage converter with power will be described.

In the above, the gate voltage of Q_O has been presumed to be V_CC +V_th. This has intended to simplify the computation and to clearly elucidate the characteristics of the circuits. Essentially, however, this voltage need not be limited to V_CC +V_th, but may be chosen at will for the convenience of design.

FIG. 29(A) shows a practicable circuit which boosts the gate voltage V_G to above the supply voltage V_CC within the chip as stated with reference to FIG. 15.

When a pulse φ_i of amplitude V_CC from an oscillator OSC included within the chip rises from 0 (zero) V to V_CC, a node 4' having been previously charged to V_CC -V_th by Q₁ ' is boosted to 2 V_CC -V_th.

In consequence, a node 4 becomes a voltage 2 (V_CC -V_th) lowered by V_th by means of Q₂ '. Subsequently, when φ_i becomes 0 V and a node 2 rises to V_CC, the node 4 is further boosted into 3 V_CC -2 V_th. Accordingly, a node 5 becomes a voltage 3 (V_CC -V_th) lowered by V_th by means of Q₂. Each of Q₂ ' and Q₂ is a kind of diode, so that when such cycles are continued a large number of times, V_G becomes a D.C. voltage of 3 (V_CC -V_th). V_G of higher voltage is produced by connecting the circuits CP1, CP2 in a larger number of stages. The reason why the two stages are comprised here, is as follows. Assuming V_CC to lower to 2.5 V and V_th to be 1 (one) V, one stage affords V_G =2 (V_CC -V_th), and hence, V_G =3 V holds. Under this condition, however, the source voltage V_L of Q_O in FIG. 15 becomes 2 V lower than V_CC. In contrast, when the two stages are disposed, V_G =4.5 V holds because of V_G =3 (V_CC -V_th). Accordingly, V_L can be equalized to V_CC, so that V_L =V_CC can be established below V_O as in FIG. 15. Conversely, however, as V_CC becomes a higher voltage, it is more of a concern that V_G may become an excess voltage which can break down the associated transistors. Therefore, some circuit for limiting V_G is required on the high voltage side of V_CC.

FIG. 30 shows an example in which V_G ≃3 (V_CC -V_th) is held as a high voltage on the low voltage side of V_CC, and besides, V_CC +2 V_th is held on the high voltage side of V_CC in order to protect the associated transistors. Here, any of the circuits thus far described, for example, the whole circuit in FIG. 16, 18, 20, 22, 24 or 26, is indicated by LM1 as the load of V_G. A protection circuit CL1 is such that, when V_G is going to exceed V_CC +2 V_th, current flows through Q₁ and Q₂, so V_G results in being fixed to V_CC +2 V_th. With the present circuit, V_CC at which CL1 operates ranges from 3 (V_CC -V_th)=V_CC +2 V_th to V_CC =5/2 V_th.

FIG. 31 shows a practicable circuit of the inverter 1 or 2 in FIG. 29(A). An output pulse φ_O is impressed on the circuit CP1 or CP2.

While the oscillator OSC can be constructed as a circuit built in the chip, FIG. 32 shows an example utilizing a back bias generator which is built in the chip in order to apply a back bias voltage V_BB to a silicon substrate. The advantage of this example is that the oscillator need not be designed anew, which is effective for reducing the area of the chip. In general, when V_L is applied to respective transistors with V_BB being 0 (zero) V, the threshold voltages V_th of the respective transistors are not normal values. Therefore, an excess current flows, or stress conditions on the transistors become severe, so the transistors can break down. In contrast, when this circuit is used, V_BB is generated upon closure of a power source, and V_L is generated substantially simultaneously, so that the operations of respective transistors are normally executed.

Next, practicable embodiments of buffer circuits will be described. As the load of the voltage converter, there is sometimes disposed a load of large capacity or of great load fluctuation. In this case, such a heavy load needs to be driven through a buffer circuit of high driving ability. In order to accomplish this, the ordinary method is to drive the load through a single transistor of high driving ability, namely, a transistor having a large width-to-length ratio W/L as shown in FIG. 33. With this method, however, the performance degrades because a voltage drop of V_th arises on the low voltage side of V_CC as shown in FIG. 34. FIG. 35 shows a practicable example of the buffer circuit which has a high driving ability without the V_th drop. When a voltage V_PP is made greater than V_L =V_th and a resistance R_P is made much higher than the equivalent "on" resistance of a transistor Q₁, the gate voltage of a transistor Q₂ becomes V_L +V_th. Accordingly, the source voltage V_L1 of Q₂ equalizes to V_L. When the W/L of Q₂ is made great, the desired buffer circuit is provided. Here, V_L becomes V_CC on the low voltage side of V_CC, so that V_PP must be at least V_CC -V_th. As a circuit therefor, the circuit shown in FIG. 29(A) is usable. Regarding connection, the node 5 of the circuit in FIG. 29(A) may be connected to the drain of Q₁ in a regulator in FIG. 35. Here, in order that the effective output impedance as viewed from the node 5 may be made sufficiently higher than the equivalent "on" resistance of Q₁ of the circuit in FIG. 35, the value of the W/L of Q₂ or the value of C_B in FIG. 29(A) or the oscillation frequency of OSC may be properly adjusted by way of example.

As to some loads, it is necessary to apply V_L to the drain of a transistor constituting a part of the load and to apply V_L +V_th to the gate thereof, so as to prevent the V_th drop and to achieve a high speed operation. FIG. 36 shows an embodiment therefor. The circuit LM₁ is, for example, the circuit in FIG. 16, and the voltage V_L1 equalizes to V_L as stated before. In addition, the gate voltage of Q₄ is V_L +2 V_th. Therefore, V_L2 becomes V_L +V_th. Here, transistors Q₆ and Q₇ serve to prevent unnecessary charges from remaining in V_L1 at the transient fluctuation of V_CC. Q₆ and Q₇ are connected into LM1 as shown in figure so as to operate at V_CC of at least V_O and at V_CC of at least V_O -V_th. Here, the ratio V/L of Q₆, Q₇ is selected to be sufficiently smaller than that of Q₂, minimize the influence of the addition of Q₆, Q₇ on V_L. It has been previously stated that Q₇ operates in the region not greater than V_O. Since Q₂ and Q₄ are in the operating states of unsaturated regions (V_GS -V_th ≧V_DS, V_GS : gate-source voltage, V_DS : drain-source voltage) in the region not greater than V_O, surplus charges are discharged to V_CC through Q₂, Q₄, and hence, Q₇ is unnecessary in principle. However, when V_CC is near V_O, the "on" resistances of Q₂, Q₄ increase unnecessarily, and it is sometimes impossible to expect the effects of these transistors. Accordingly, Q₇ is added, whereby stable values of V_L1 can be obtained in a wide range from the region (V_O -V_th) where V_CC is not greater than V_O, to the region where V_CC is greater than V_O and where the converter is normally operating.

The function of Q₅ is that, when V_L1 is going to fluctuate negatively relative to V_L2, current flows to Q₅ so as to keep the difference of V_L2 and V_L1 constant. In addition, in the present embodiment, the example of V_L and V_L +V_th has been stated. However, when the pairs of Q₁, Q₂ or the pairs of Q₃, Q₄ are connected in cascade, a voltage whose difference from V_L1 becomes an integral multiple of V_th can be generated.

A circuit shown in FIG. 37 is another buffer circuit which is connected to the output stage of the circuit of FIG. 35 or 36 in order to further enhance the driving ability of the buffer circuit of FIG. 35 or 36. By connecting such a buffer circuit of higher driving ability, a large load capacity can be driven. First, V_L1 becomes V_L1 +2 V_th and V_L1 +V_th at respective nodes 4 and 2. Eventually, however, it is brought into V_DP being the level of V_L1 at a node 5 by Q₄. Problematic here is the load driving ability of Q₄ which serves to charge a large capacitance C_D in the load LC1 at high speed. In order to enhance the ability, the node 2 being the gate of Q₄ needs to be boosted in a time zone for charging the load. Transistors therefor are Q₆ -Q₁1, and capacitors are C₁ and C₂. A node 6 discharged by Q₁3 owing to the "on" state of φ₂ is charged by Q₁2 and Q₄ when the next φ₁ is in the "on" state. At this time, the node 2 being at V_L1 +V_th and a node 3 being at V_L1 are boosted by the "on" state of φ₁. Consequently, the conductances of Q₁0, Q₁1 increase, so that the boosted voltage of the node 2 is discharged to the level of V_L1 +V_th by Q₁0, Q₁1. Here, when the boosting time is made longer than the charging time of C_D based on Q₄, Q₁2, the capacitor C_D is charged rapidly. The transistor Q₆ cuts off the nodes 3 and 1 when the node 3 is boosted by φ₁. When φ₂ is "on", Q₇ -Q₉ turn "off" subject to the condition of V_L1 ≦3V_th, so that Q₁1 has its gate rendered below V_th to turn "off". Accordingly, no current flows through Q₃, Q₁0 and Q₁1, so that the power consumption can be rendered low. In addition, in order to reduce to power consumption in the case of V_L1 >3 V_th, the "on" resistance of Q₆ may be increased to lower current. The voltage of the node 3 at this time becomes a stable value of approximately 3 V_th. Thus, the boosting characteristic of the node 3 is also stabilized, with the result that the operation of the whole circuit can be stabilized.

Here, since the sources and gates of Q₇ and Q₁0 are connected in common, the conditions of biasing the gates are quite equal. Accordingly, when ##EQU7## is held in advance, the boosting characteristics of the nodes 2, 3 can be quite equal, so the circuit design can be facilitated advantageously. That is, one merit of the present embodiment consists in that the boosting characteristic of the node 2 can be automatically controlled with the boosting characteristic of the node 3. In this way, the D.C. path from the node 2 to V_SS in the case of performing no boosting can be relieved, and it becomes possible to lower the power consumption.

Here, Q₅ has the function of discharging the surplus charges of the node 2 when Q₁0 is "off".

As regards the embodiment of FIG. 37, various modifications can be considered. While the drain of Q₆ in FIG. 37 is connected to V_L1 in order to stabilize the boosting characteristics of the nodes 2, 3 to the utmost, it can also be connected to V_CC so as to relieve a burden on V_L1. Likewise, while Q₁0 subject to the same operating condition as that of Q₇ is disposed in order to stabilize the boosting characteristics of the nodes 2, 3, it may well be removed into an arrangement in which the nodes 2 and 9 are directly connected, with the source of Q₇ and the node 9/disconnected. Since, in this case, the relationship of Q₉ and Q₁1 is in the aforementioned relationship of Q₇ and Q₁0, the boosting characteristics can be similarly designed, and the occupying area of the circuit can be effectively reduced. Further, the 3-stage connection arrangement of Q₇, Q₈ and Q₉ is employed here. This is a consideration for efficiently forming the circuit in a small area by utilizing a capacitance Chd 2 (for example, the capacitance between the gate of a MOST and an inversion layer formed between the source and drain thereof, known from ISSCC 72 Dig. of Tech. Papers, p. 14, etc.) for the reduction of the power consumption described above. That is, in order to use the inversion layer capacitance, the gate voltage to be applied needs to be higher by at least V_th than the source and drain. Accordingly, in case of forming C₂ by the use of a MOST of low V_th or an ordinary capacitor, it is also possible to reduce the connection number of Q₇ -Q₉ to two or one.

The buffer circuit as shown in FIG. 37 is indispensable especially to the LSI systems as shown in FIGS. 1 and 2. In general, the voltage converter for generating V_L in FIG. 1 or 2 is desired to have an especially high ability of supplying current because the circuit current in the circuit A, A' or B flows toward the ground. Accordingly, when the whole circuit including the circuit of FIG. 37 thus far described is regarded as the voltage converter of FIG. 1 or 2, it is applicable to general LSIs.

With the embodiments stated above, when the actual circuit of FIG. 18 which is diode-connected as shown in FIG. 27 is operated at V_CC of or above V_O as shown in FIG. 17, current flows through Q₁ '-Q_S ' (FIG. 27) to increase the power consumption. This increase of the power consumption poses a problem in case of intending to back up the LIS power source, namely, the externally applied supply voltage with a battery. More specifically, in an apparatus wherein the ordinary external power source is backed up by a battery when turned "off"; when the power consumption of the LSI itself is high, the period of time for which the power source is backed up is limited because the current capacity of the battery is small. Therefore, with a measure wherein V_CC to be applied by the battery is set at below V_O during the time interval during which the battery is operated for backup, no current flows through Q₁ '-Q_S ', and hence, the period of time for which the power source can be backed up can be extended to that extent. Alternatively, the number of stages of Q₁ '-Q_S ' (FIG. 27) can be determined so as to establish V_O which is greater than V_CC being the battery supply voltage in the case of the backup.

The supply voltage V_CC in the ordinary operation can be selected at V_CC <V_O besides at V_CC >V_O. Since this permits no current to flow through Q₁ '-Q_S ' under the ordinary V_CC condition, the power consumption can be lowered. Another merit is that design is facilitated because the circuit can be designed while avoiding a region where the relation of V_CC and V_L becomes a polygonal line. More specifically, when the polygonal region is used, an imbalance of characteristics concerning V_CC arises between a circuit directly employing V_CC and a part of a certain circuit employing V_L by way of example, so that the operation sometimes becomes unstable. When V_CC <V_O holds, this drawback can be eliminated.

In the above, the practicable embodiments have been described in which the voltage converters are constructed of MOS transistors. These are examples which chiefly employ MOS transistors of positive threshold voltages V_th, namely, of the enhancement mode. Needless to say, however, it is also possible to employ a MOS transistor of negative V_th, namely, of the depletion mode as disclosed in FIG. 16 of Japanese Patent Application No. 56-168,698. For example, in the embodiment of FIG. 16, in order to establish V_L =V_CC in the region of V_CC ≦V_O as illustrated in the characteristic of FIG. 15, the gate voltage of Q_O needs to be V_G ≧V_CC +V_th(O), and it has been stated that the circuit of FIG. 29(A) may be used as the V_G generator therefor. In this regard, the circuit can be further simplified by employing the MOS transistor of the depletion mode. FIG. 39 shows such a practicable embodiment. It differs from the circuit of FIG. 16 in that Q_O is replaced with the depletion mode MOS transistor Q_O ', the gate of which is connected to the terminal 2. With this measure, since the V'_th(O) of Q_O ' is negative, Q_O ' is in the "on" state at all times, and the desired characteristic illustrated in FIG. 15 can be realized without employing the V_G generator as shown in FIG. 29(A). With the present embodiment, not only the circuit arrangement can be simplified as stated above, but also the merit of attaining a stable characteristic is achieved because current I(Q_O ') to flow through Q_O ' becomes a constant current determined by β'(O) (channel conductance) and V'_th(O) (threshold voltage) as ##EQU8## Although the present embodiment has exemplified FIG. 16, it is applicable as it is by substituting Q_O ' for Q_O in any other embodiment and connecting its gate to the terminal 2 as in the present embodiment.

FIG. 40 shows an embodiment in which a buffer circuit is constructed using a single depletion-mode MOS transistor, while FIG. 41 shows the characteristic thereof. Although the present embodiment is the same in the circuit arrangement as the foregoing embodiment of FIG. 33, it differs in that the MOS transistor is changed from the enhancement mode into the depletion mode. As shown in FIG. 41, the output V_L' of the present buffer circuit bends from a point P at which the difference of V_CC and V_L equalizes to the absolute value |V_thD | of the threshold voltage V_thD of the MOS transistor, and it thereafter becomes a voltage which is higher than V_L by |V_thD |. Accordingly, V_L may be set lower than a desired value by |V_thD |. The present embodiment has a simple circuit arrangement, and can meritoriously eliminate the problem, as in the characteristic of the embodiment of FIG. 33 illustrated in FIG. 34, that only the output lower than V_CC by V_th can be produced in the range of V_CC ≦V_O.

As set forth above, the present invention can provide, in an integrated circuit having small geometry devices, an integrated circuit which has a wide operating margin even against the fluctuations of an external supply voltage in an ordinary operation and which can apply a sufficient aging voltage.

It is to be understood that the above-described arrangements are simply illustrative of the application of the principles of this invention. Numerous other arrangements may be readily devised by those skilled in the art which embody the principles of the invention and fall within its spirit and scope.

Claims

1. A semiconductor integrated circuit contained within a chip having a regulated voltage comprising:

circuit means contained within the semiconductor integrated circuit which has an input to which is coupled the regulated voltage for powering said circuit means;

means for producing the regulated voltage having an input adapted to be coupled to a means for supplying an external supply voltage, and an output which is coupled to the input of the circuit means to which the regulated voltage is applied, the regulated voltage being a function of the external supply voltage whose rate of change with respect to change in the external supply voltage is not constant when external supply voltage is changed from a value thereof in a predetermined ordinary operation range of the semiconductor integrated circuit to a different value within a range for performing an aging test on said semiconductor integrated circuit, said means for producing the regulated voltage comprising a first transistor having an input terminal to which a reference voltage is applied from a reference voltage generator producing a reference voltage and an output which is the regulated voltage, the output also being applied to a second transistor coupled to a reference potential which applies a negative feedback to at least one additional transistor coupled between the output of the first transistor and the second transistor; and

the regulated voltage provided during the aging test approximately equalizing stress conditions of transistors of small geometries receiving the regulated voltage to stress conditions of transistors of large geometries receiving the external voltage.

2. A semiconductor integrated circuit according to claim 1, wherein the rate of change of the output voltage of said means for producing the regulated voltage is less when said external supply voltage changes within the ordinary operation range than the rate of change of the output voltage when the external supply voltage changes within the range for performing said aging test.

3. A semiconductor integrated circuit according to claim 1, wherein the change of the output voltage of said means for producing the regulated voltage follows the change of said external supply voltage when said external supply voltage changes within the ordinary operation range.

4. A semiconductor integrated circuit according to claim 1, 2 or 3, wherein said means for producing the regulated voltage changes its output voltage so as to reach unequal aging voltages in succession in accordance with the change of said external supply voltage.

5. A semiconductor integrated circuit according to claim 1, 2 or 3, wherein said means for producing the regulated voltage includes means to prevent its output voltage from exceeding a breakdown voltage of the internal circuit when said output voltage of said means for producing the regulated voltage has exceeded a predetermined aging voltage.

6. A semiconductor integrated circuit according to claim 5, wherein said means for producing the regulated voltage includes means to negatively change its output voltage in correspondence with a rise of said external supply voltage when said output voltage of said means for producing the regulated means voltage has exceeded the predetermined aging voltage.

7. A semiconductor integrated circuit according to claim 5, wherein said means for producing the regulated voltage includes means to prevent its output voltage from exceeding a breakdown voltage of the circuit means when said output voltage of said means for producing the regulated voltage has exceeded a predetermined aging voltage.

8. A semiconductor integrated circuit according to claim 5, wherein said means for producing the regulated voltage includes means to negatively change its output voltage in correspondence with a rise of said external supply voltage when said output voltage of said means for producing the regulated voltage has exceeded the predetermined aging voltage.

9. A semiconductor integrated circuit in accordance with claim 1 wherein:

the reference voltage producted produced by the reference voltage generator is zero when operation of the circuit means is in the ordinary range and increases in a step function from a first potential of zero to a first level at a lowest voltage of operation of the aging test and increase increases during higher potentials of the external voltage during the conducting of the aging test.

10. A semiconductor integrated circuit in accordance with claim 9 wherein:

the first transistor is a FET having a gate coupled to the reference voltage, one of a source and a drain is coupled to a reference potential and the other of the source and drain is the output;

the second transistor is a FET having one of a source or drain coupled to the output and the other of the source and drain is coupled to a reference potential; and

each additional transistor having a gate coupled to one of a source and drain with the source and drain of each of the one or more additional transistors being coupled in a series circuit coupled between the output of the first transistor and a gate of the first transistor.

11. A semiconductor integrated circuit in accordance with claim 1 wherein:

the regulated voltage varies as a linear function of the external supply voltage during a predetermined ordinary operation range.

12. A semiconductor integrated circuit in accordance with claim 11 wherein:

the regulated voltage in the range for performing an aging test has a slope for changes in the external supply voltage which is less than a slope for changes in the external supply voltage in the range of ordinary operation.

13. A semiconductor integrated circuit comprising:

an input terminal for receiving an external supply voltage; and

means coupled to said input terminal for lowering said supply voltage to provide an output voltage at a reduced power supply terminal for providing power at a reduced voltage level to at least one predetermined circuit coupled to said reduced power supply terminal, said lowering means comprising:

means for providing a first output voltage signal which is a function of the external supply voltage and having a first predetermined rate of change with respect to change in the external supply voltage for change in said supply voltage from a first predetermined supply voltage greater than zero to a second predetermined supply voltage with the first and second predetermined voltages defining a first range for operating the circuit in a predetermined ordinary operating range;

means for providing a second output voltage signal which is a function of the external supply voltage and having a second predetermined rate of change different than that of said first predetermined rate of change with respect to change in the external supply voltage for change in said supply voltage from said second predetermined supply voltage to a third predetermined supply voltage greater than said second predetermined supply voltage with the second and third predetermined voltages defining a second range for performing an aging test on the circuit;

means for providing a third output voltage signal which is a function of the external supply voltage and having a third predetermined rate of change different than that of said second predetermined rate of change with respect to change in the external supply voltage for change in said supply voltage from said third predetermined supply voltage to a fourth predetermined supply voltage greater than said third predetermined supply voltage with the third and fourth predetermined voltages defining a third range for performing an aging test on the circuit; and

said second or third output voltage signals provided during the aging test approximately equalizing stress conditions of transistors of small geometries receiving said second or third output voltage signal to stress conditions of transistors of large geometries receiving said external supply voltage.

14. A semiconductor integrated circuit according to claim 13, wherein said first predetermined supply voltage is zero.

15. A semiconductor integrated circuit according to claim 13, wherein said second and third rates of change are positive, and said second rate of change is less than said third rate of change.

16. A semiconductor integrated circuit according to claim 13, wherein said second rate of change is positive and said third rate of change is negative.

17. A semiconductor integrated circuit according to claim 13, wherein said second rate of change is zero and said third rate of change is positive.

18. A semiconductor integrated circuit according to claim 13, wherein said second rate of change is zero and said third rate of change is negative.

19. A semiconductor integrated circuit according to claim 13, wherein said lowering means further comprises means for providing a fourth output voltage signal which is a function of the external supply voltage and having a fourth predetermined rate of change different from that of said third predetermined rate of change with respect to change in the external supply voltage for change in said supply voltage between said fourth predetermined supply voltage and a fifth predetermined supply voltage greater than said fourth predetermined supply voltage with the fourth and fifth predetermined voltages defining a fourth range for performing an aging test on the circuit.

20. A semiconductor integrated circuit according to claim 19, wherein said second, third and fourth rates of change are positive, and wherein said second rate of change is less than said third rate of change and said third rate of change is less than said fourth rate of change.

21. A semiconductor integrated device according to claim 19, wherein said second, third and fourth rates of change are positive, and wherein said second rate of change is greater than said third rate of change and said third rate of change is greater then said fourth rate of change.

22. A semiconductor integrated circuit according to claim 19, wherein said second rate of change is zero, and said third and fourth rates of change are positive, with said third rate of change being greater than said fourth rate of change.

23. A semiconductor integrated circuit according to claim 19, wherein said second rate of change is zero and said third and fourth rates of change are positive, with said third rate of change being less than said fourth rate of change.

24. A semiconductor integrated circuit contained within a chip having a regulated voltage comprising:

circuit means contained within the semiconductor integrated circuit which has an input to which is coupled the regulated voltage for powering said circuit means; and

means for producing the regulated voltage having an input adapted to be coupled to a means for supplying an external supply voltage, and an output which is coupled to the input of the circuit means to which the regulated voltage is applied, the regulated voltage being a function of the external supply voltage whose rate of change with respect to change in the external supply voltage is not constant when said external supply voltage is changed from a value thereof in a predetermined ordinary operation range of the semiconductor integrated circuit to a higher value within a range for performing an aging test on said semiconductor integrated circuit, the regulated voltage provided during the aging test approximately equalizing stress conditions of transistors of small geometries receiving the regulated voltage to stress conditions of transistors of large geometries receiving the external supply voltage, the means for producing the regulated voltage comprising a reference voltage generator coupled to the external supply voltage for producing a control voltage which varies as a function of the external supply voltage and a transistor having a current conduction path coupled between the external supply voltage and a reference potential and a control terminal to which is applied the control voltage, the output being coupled to the current conduction path at a point remote from the reference potential.

25. A semiconductor integrated circuit within a chip having a regulated voltage in accordance with claim 24, wherein the means for producing a regulated voltage produces a rate of change during normal operation and the conducting of an aging test which is not constant.

26. A semiconductor integrated circuit contained within a chip having a regulated voltage comprising:

circuit means contained within the semiconductor integrated circuit which has an input to which is coupled the regulated voltage for powering said circuit means; and

means for producing the regulated voltage having an input adapted to be coupled to a means for supplying an external supply voltage, and an output which is coupled to the input of the circuit means to which the regulated voltage is supplied, the regulated voltage being a function of the external supply voltage whose rate of change with respect to change in the external supply voltage has a first rate of change between an external voltage varying between 0 volts and a first predetermined magnitude and a second non-constant rate of change from the first predetermined magnitude to a greater magnitude, the second non-constant rate of change including a predetermined ordinary operation range for the circuit means and a higher range for conducting an aging test, the rate of change for the ordinary operation range being different from the rate of change for conducting an aging test, and the rate of increase of the regulated voltage as a function of the external voltage in the range between 0 volts and the first predetermined magnitude being greater than the average rate of increase between the first predetermined magnitude and the greater magnitude of the second non-constant rate of change, the regulated voltage provided during the aging test approximately equalizing stress conditions of transistors of small geometries receiving the regulated voltage to stress conditions of transistors of large geometries receiving the external supply voltage.

27. A semiconductor integrated circuit as in claim 26, wherein the rate of change for the ordinary operation range is less than the rate of change for the conducting of the aging test.