Communication system

Abstract

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into, low and high, frequency band components which are designated as a first and a second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

Description

transmission32
|A(16)|²=(A₁+A₂)²+(B₁+B₂)²=4A_T0²+4A_T0²=8_T0²
|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greater amplitude and a four times greater transmitting energy than those needed for the 4 PSK signal. A modified 16 QAM signal according to the present invention will not be demodulated by a common receiver designed for symmetrical, equally distanced signal point QAM. However, it can be demodulated with the second receiver 33 when two threshold A₁ and A₂ are predetermined to appropriate values. At FIG. 10, the minimum distance between two signal points in the first segment of the signal point group 91 is A₁ and A₂/2A₁ is established as compared with the distance 2A₁ of 4 PSK. Then, as A₁=A₂, the distance becomes ½. This explains that the signal receiving sensitivity has to be two times greater for the same error rate and four times greater for the same signal level. For having a four times greater value of sensitivity, the radius r₂ of the antenna 32 of the second receiver 33 has to be two times greater than the radius r₁ of the antenna 22 of the first receiver 23 thus satisfying r₂=2r₁. For example, the antenna 32 of the second receiver 33 is 60 cm diameter when the antenna 22 of the first receiver 23 is 30 cm. In this manner, the second data stream representing the high frequency component of an HDTV will be carried on a signal channel and demodulated successfully. As the second receiver 33 intercepts the second data stream or a higher data signal, its owner can enjoy a return of high investment. Hence, the second receiver 33 of a high price may be accepted. As the minimum energy for transmission of 4 PSK data is predetermined, the ratio n₁₆ of modified 16 APSK transmitting energy to 4 PSK transmitting energy will be calculated to the antenna radius r₂ of the second receiver 33 using a ratio between A₁ and A₂ shown in FIG. 10.

In particular, n₁₆ is expressed by ((A₁+A₂)/A₁)² which is the minimum energy for transmission of 4 PSK data. As the signal point distance suited for modified 16 QAM interception is A₂, the signal point distance for 4 PSK interception is 2A₁, and the signal point distance ratio is A₂/2A₁, the antenna radius r₂ is determined as shown in FIG. 11, in which the curve 101 represents the relation between the transmitting energy ratio n₁₆ and the radius r₂ of the antenna 22 of the second receiver 23.

Also, the point 102 indicates transmission of common 16 QAM at the equal distance signal state mode where the transmitting energy is nine times greater and thus will no more be practical. As apparent from the graph of FIG. 11, the antenna radius r₂ of the second receiver 23 cannot be reduced further even if n₁₆ is increased more than 5 times.

The transmitting energy at the satellite is limited to a small value and thus, n₁₆ preferably stays not more than 5 times the value, as denoted by the hatching of FIG. 11. The point 104 within the hatching area 103 indicates, for example, that the antenna radius r₂ of a two times greater value is matched with a 4×value of the transmitting energy. Also, the point 105 represents that the transmission energy should be doubled when r₂ is about 5×greater. Those values are all within a feasible range.

The value of n₁₆ not greater than 5×value is expressed using A₁ and A₂ as:
n₁₆=((A₁+A₂)/A₁)²≦5
Hence, A₂≦1.23A₁.

If the distance between any two signal point group segments shown in FIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) and A(16)−A(4) are proportional to A₁ and A₂ respectively. Hence, (A(16))²≦5(A(14))² is established.

The action of a modified 64 ASPK transmission will be described as the third receiver 43 can perform 64-state QAM demodulation.

FIG. 12 is a vector diagram in which each signal point group segment contains 16 signal points as compared with 4 signal points of FIG. 10. The first signal point group segment 91 in FIG. 12 has a 4×4 matrix of 16 signal points allocated at equal intervals including the point 170. For providing compatibility with 4 PSK, A₁≧A_T0 has to be satisfied. If the radius of the antenna 42 of the third receiver 43 is r₃ and the transmitting energy is n₆₄, the equation is expressed as:
r₃²={6²/(n−1)}r₁²

This relation between r₃ and n of a 64 QAM signal is also shown in the graphic representation of FIG. 13.

It is understood that the signal point assignment shown in FIG. 12 allows the second receiver 33 to demodulate only two-bit patterns of 4 PSK data. Hence, it is desired for having compatibility between
This
Therefore, the antenna 32 of the second receiver 33 has to be a little bit increased in radius as denoted by the curve 213.

As understood, while the first and second data streams are transmitted

The various data for demodulation including A₁ and A₂ or TH₁₆, and the value m for multiple-bit modulation are also transmitted from the transmitter 1 as carried in the first data stream. The demodulation controller 231 may be arranged for recovering such demodulation data through statistic process of the received signal.

A way of determining the shift factor A₁/A₂ will be described with reference to FIG. 26. A change of the shift factor A₁/A₂ causes a change of the threshold value. Increase of a difference of a value of A₁/A₂ set at the receiver side from a value of A₁/A₂ set at the transmitter side will increase the error rate. Referring to FIG. 26, the demodulated signal from the second data stream reproducing unit 233 may be fed back to the demodulation controller 231 to change the shift factor A₁/A₂ in a direction to increase the error rate. By this arrangement, the third receiver 43 may not demodulate the shift factor A₁/A₂, so that the circuit construction can be simplified. Further, the transmitter may not transmit the shift factor A₁/A₂, so that the transmission capacity can be increased. This technique can be applied also to the second receiver 33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for the C-CDM signal points, wherein signal points are added by shifting in the polar coordinate direction (r, θ). The previously described C-CDM is characterized in that the signal points are shifted in the rectangular coordinate direction, i.e. XY direction; therefore it is referred to as rectangular coordinate system C-CDM. Meanwhile, this C-CDM characterized by the shifting of signal points in the polar coordinate direction, i.e. r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein four signal points are added by shifting each of 4 QPSK signals in the radius r direction of the polar coordinate system. In this manner, the APSK of polar coordinate system C-CDM having 8 signal points is obtained from the QPSK as shown in FIG. 25(a). As the pole is shifted in the polar coordinate system to add signal points in this APSK, it is referred to as shifted pole-APSK, i.e SP-APSK in the abbreviated form. In this case, coordinate value of the newly added four QPSK signals 85 are specified by using a shift factor S₁ as shown in FIG. 139. Namely, 8PS-APSK signal points
This calculation is equivalent to that of TH₁₆ but its resultant distance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A₃, the distance between the first 181 and the second sub segment 182 is expressed by (A₂−2A₃). Then, the average distance is (A₂−2A₃)/(A₁+A₂) which is designated as d₆₄. When d₆₄ is smaller than T₂ which represents the signal point discrimination capability of the second receiver 33, any two signal points in the segment will hardly be distinguished from each other. This judgement is executed at Step
TH2₆₄=(A₁+A₂/2)/(A₁+A₂)
and
TH3₆₄=(A₁+A₂−A₃/2)/(A₁+A₂)

Through AM demodulation of a phase detected signal using the three threshold values, the third data stream can be reproduced like the first and second data stream explained with FIG. 21. The third data stream contains e.g. four signal points 201, 202, 203, 204 at the first sub segment 181 shown in FIG. 23 which represent 4 values of two-bit pattern. Hence, six digits or modified 64 QAM signals can be demodulated.

The demodulation controller 231 detects the value m, A₁, A₂, and A₃ from the demodulation data contained in the first data stream demodulated at the first data stream reproducing unit 232 and calculates the three threshold values TH1₆₄, TH2₆₄, and TH3₆₄ which are then fed to the first 136 and the second discrimination/reproduction circuit 137 so that the modified 64 QAM signal is demodulated with certainty. Also, if the demodulation data have been scrambled, the modified 64 QAM signal can be demodulated only with a specific or subscriber receiver. FIG. 28 is a flow chart showing the action of the demodulation controller 231 for modified 64 QAM signals. The difference from the flow chart for demodulation of 16 QAM shown in FIG. 24 will be explained. The procedure moves from Step 304 to Step 320 where it is examined where m=32 or not. If m=32, demodulation of 32 QAM signals is executed at Step 322. If not, the procedure moves to Step 321 where it is examined whether m=64 or not. If yes, A₃ is examined at Step 323. If A₃ is smaller than a predetermined value, the procedure moves to Step 305 and the same sequence as of FIG. 24 is implemented. If it is judged at Step 323 that A₃ is not smaller than the predetermined value, the procedure goes to Step 324 where the threshold values are calculated. At Step 325, the calculated threshold values are fed to the first and second discrimination/reproduction circuits and at Step 326, the demodulation of the modified 64 QAM signal is carried out. Then, the first, second, and third data streams are reproduced at Step 327. At Step 328, the error rate is examined. If the error rate is high, the procedure moves to Step 305 where the 16 QAM demodulation is repeated and if low, the demodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of a satisfactory demodulating procedure will now be described. The scope of the present invention includes reproduction of the first data stream of a modified 16 or 64 QAM signal with the use of a 4 PSK receiver. However, a common 4 PSK receiver rarely reconstructs carrier waves, thus failing to perform a correct demodulation. For compensation, some arrangements are necessary at both the transmitter and receiver sides.

Two techniques for the compensation are provided according to the present invention. A first technique relates to transmission of signal points aligned at angles of (2n−1)π/4 at intervals of a given time. A second technique offers transmission of signal points arranged at intervals of an angle of nπ/8.

According to the first technique, the eight signal points including 83 and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown in FIG. 38. In action, at least one of the eight signal points is transmitted during sync time slot periods 452, 453, 454, 455 arranged at equal intervals of a time in a time slot gap 451 shown in the time chart of FIG. 38. Any desired signal points are transmitted during the other time slots. The transmitter 1 is also arranged to assign a data for the time slot interval to the sync timing data region 499 of a sync data block, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detail referring to FIG. 41. The time slot group 451 containing the sync time slots 452, 453, 454, 455 represents a unit data stream or block 491 carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of a given time determined by the time slot interval or sync timing data. Hence, when the arrangement of the sync time slots is detected, reproduction of carrier waves will be executed slot by slot through extracting the sync timing data from their respective time slots. Such a sync timing data S is contained in a sync block 493 accompanied at the front end of a data frame 492, which consists of a number of the sync time slots denoted by the hatching in FIG. 41. Accordingly, the data to be extracted for carrier wave reproduction are increased, thus allowing the 4 PSK receiver to reproduce desired carrier waves at higher accuracy and efficiency.

The sync block 493 comprises sync data regions 496, 497, 498, - - - containing sync data S1, S2, S3, - - - respectively which include unique words and demodulation data. The phase sync signal assignment region 499 is accompanied at the end of the sync block 493, which holds a data of I_T including information about interval arrangement and assignment of the sync time slots.

The signal point data in the phase sync time slot has a particular phase and can thus be reproduced by the 4 PSK receiver. Accordingly, I_T in the phase sync signal assignment region 499 can be retrieved without error thus ensuring the reproduction of carrier waves

The other signal points 84a and 86a are also shifted to two points 84 and 86 respectively.

If the error rate of the first data stream is Pe₁, it is obtained from: $\begin{array}{c}\mathrm{Pe1}-16=\frac{1}{4}\mathrm{erfc}\left(\frac{n\delta }{\sqrt{2\sigma }}\right)+\mathrm{erfc}\left(\frac{3\delta }{\sqrt{2\sigma }}\right)\\ =\frac{1}{8}\mathrm{erfc}\left(\frac{n\sqrt{\rho }}{\sqrt{9+n^2}}\right)\end{array}$
Also, the error rate Pe₂ of the second data stream is obtained from: $\begin{array}{c}\mathrm{Pe2}-16=\frac{1}{2}\mathrm{erfc}\left(\frac{\frac{3-n}{2}\delta }{\sqrt{2\sigma }}\right)\\ =\frac{1}{4}\mathrm{erfc}\left(\frac{3-n}{2\sqrt{9+n^2}}\sqrt{\rho }\right)\end{array}$

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is a vector diagram of a 36 SRQAM signal in which the distance between any two 36 QAM signal points is
Hence, if the 32 SRQAM signal is selected, the shift n is determined by:
1<n<5
Also, if the 16 SRQAM signal is employed, n is determined by:
1<n<3

In the SRQAM mode signal terrestrial broadcast service in which the first and second data levels are created by shifting corresponding signal points as shown in FIGS. 99 and 100, the advantage of the present invention will be given when the shift n is a 16, 32, or 64 SRQAM signal is more than 1.0.

In the above embodiments, the low and high frequency band components of a video signal are transmitted as the first and second data streams. However, the transmitted signal the
Hence, the multi-level signal transmission system of the embodiment is based on L_O>L. The embodiment is however not limited to L_O>L and L=L_O will be employed temporarily or permanently depending on the requirements of design, condition, and setting. FIGS. 68(a) and 68(b) show constellation for 8-value VSB.

The two signal point groups are assigned one-bit patterns of the first data stream D₁, as shown in FIG. 59(a). More particularly, a bit 0 of binary system is assigned to the first signal point group 725 and another bit 1 to the second signal point group 726. Then, a one-bit pattern of the second data stream D₂ is assigned to each signal point. For example, the two signal points 721, 723 are assigned D₂=0 and the other two signal points 722 and 724 are assigned D₂=1. Those are thus expressed by two bits per symbol.

The multi-level signal transmission of the present invention can be implemented in an ASK mode with the use of the foregoing signal point assignment. The system of the present invention works in the same manner as of a conventional equal signal point distance technique when the signal to noise ratio or C/N rate is high. If the C/N rate becomes low and no data can be reproduced by the conventional technique, the present system ensures reproduction of the first data stream D₁ but not the second data stream D₂. In more detail, the state at a low C/N is shown in FIG. 60—the signal allocation diagram for ASK of 4-VSB. The signal points transmitted are displaced by a Gaussian distribution to ranges 721a, 722a, 723a, 724a respectively at the receiver side due to noise and transmission distortion. Therefore, the distinction between the two signals 721 and 722 by the slice level 2 or two signals 723 and 724 by the slice level 4 will hardly be executed. In other words, the error rate in the second data stream D₂ will be increased. As apparent from FIG. 60, the two signal points 721, 722 are easily distinguished from the other two signal points 723, 724. The distinction between the two signal point groups 725 and 726 can thus be carried out with ease. As the result, the first data stream D₁ will be reproduced at a low error rate.

Accordingly, the two different level data D₁ and D₂ can be transmitted simultaneously. More particularly, both the first and second data streams D₁ and D₂ of a given signal transmitted through the multi-level transmission system can be reproduced at the area where the C/N rate is high and the first data stream D₁ only can be reproduced in the area where the C/N rate is low.

FIG. 61 is a block diagram of a transmitter 741 in which an input unit 742 comprises a first data stream input 743 and a second data stream input 744. A carrier wave from a carrier generator 64 is amplitude modulated by a multiplier 746 to generate 4- or 8-ASK signal as shown in FIG. 62(a) using an input signal fed across a processor 745 from the input unit 742. The modulated 4- or 8-ASK signal is then band limited by a band-pass filter 747 to a vestigial side band having a side band with a slight residual carrier as shown in FIG. 62(b)—an ASK signal of e.g. VSB mode which is then delivered from an output unit 748.

The waveform of the ASK signal after filtering will now be examined. FIG. 62(a) shows a frequency spectrum of the ASK modulated signal in which two sidebands are provided on both sides of the carrier frequency band. One of the two sidebands is eliminated with the filter .level. Signallevel levels, a low frequency band D_1-1, a medium-low frequency band D_1-2, and a high-medium-low frequency band D₂, video signals, and fed to an input section.

In a first data stream input 743, D_1-1 signal is ECC encoded with high code gain and D_1-2 signal is ECC coded with normal code gain. A TDM 743 743c performs time division multiplexing of D_1-1 and D_1-2 signals to produce a D₁ signal, which is then fed to a D₁ serial to parallel converter 791d in a modulator 852a. The D₁ signal consists of n pieces of parallel data, which are inputted into first inputs of n pieces of C-CDM modulator 4a, 4b, - - - respectively.

On the other hand, the high frequency band signal D₂ is fed into a second data stream input 744 of the input section 742, in which D₂ signal is ECC (Error Correction Code) encoded in an ECC 744a and then Trellis encoded in A Trellis encoder 744b. Thereafter, the D₂ signal is supplied to a D₂ serial to parallel converter 791b of the modulator 852a and converted into n pieces of parallel data, which are inputted and second inputs of the n pieces of C-CDM modulator 4a, 4b - - - respectively.

The C-CDM modulators 4a, 4b, 4c - - - respectively produces produce 16 SRQAM signal on the basis of D₁ data of the first data stream input and D₂ data of the second data stream input. These n pieces of C-CDM modulator respectively has have a carrier different from each other. As shown in FIG. 124, carriers 794a, 794b, 794c, - - - are arrayed on the frequency axis 793 so that adjacent two carriers are 90°-out-of-phase with each other. Thus C-CDM modulated n pieces of modulated signal are fed into the inverse FFT circuit 40 and mapped from the frequency axis dimension 793 to the time axis dimension 790 799. Thus, time signals 796a, 796b - - - , having an effective symbol length ts, are produced. There is provided a guard interval zone 797a of Tg seconds between the effective symbol time zones 796a and 796b, in order to reduce multipath obstruction. FIG. 129 is a graph showing a relationship between time axis and signal level. The guard time Tg of the guard interval band 797a is determined by taking account of multipath affection and usage of signal. By setting the guard time Tg longer than the multipath affected time, e.g. TV ghost, modulated signals from the inverse FFT circuit 40 are converted by a parallel to serial converter 4e into one signal and, then, transmitted from a transmitting circuit 5 as an RF signal.

Next, an action of a receiver 43 will be described. A received signal, shown as time-base symbol signal 796e of FIG. 124, is fed into an input section 24 of FIG. 123. Then, the received signal is converted into a digital signal in a demodulator 852b and further changed into Fourier coefficients in a an FFT 40a. Thus, the signal is mapped from the time axis 799 to the frequency axis 793a as shown in FIG. 124. That is, the time-base symbol signal is converted into frequency-base carriers 794a, 794b, - - - . As these carriers are in quadrature relationship with each other, it is possible to separate respective modulated signals. FIG. 125(b) shows thus demodulated 16 SRQAM signal, which is then fed to respective C-CDM demodulators 45a, 45b, - - - of a C-CDM demodulator 45, in which demodulated 16 SRQAM signal is demodulated into multi-level sub signals D₁, D₂. These sub signals D₁ and D₂ are further demodulated by a D₁ parallel to serial converter 852a and a D₂ parallel to serial converter 852b into original D₁ and D₂ signals.

Since the signal transmission system is of C-CDM multi-level shown in 125(b), both D₁ and D₂ signals will be demodulated under better receiving condition but only D₁ signal will be demodulated under worse, e.g. low C/N rate, receiving condition. Demodulated D₁ signal is demodulated in an output section 757. As the D_1-1 signal has higher ECC code gain as compared with the D_1-2 signal, an error signal of the D_1-1 signal is reproduced even under worse receiving condition.

The D_1-1 signal is converted by a 1-1 video decoder 402c into a low frequency band signal and outputted as an LDTV, and the D_1-2 signal is converted by a 1-2 video decoder 402d into a medium frequency band signal and outputted as EDTV.

The D₂ signal is Trellis decoded by a Trellis decoder 759b and converted by a second video decoder 402b into a high frequency band signal and outputted as an HDTV signal. Namely, an LDTV signal is outputted in case of the low frequency band signal only. An EDTV signal of a wide NTSC grade is outputted if the medium frequency band signal is added to the low frequency band signal, and an HDTV signal is produced by adding low, medium, and high frequency band signals. As well as the previous embodiment, a TV signal having a picture quality depending on a receiving C/N rate can be received. Thus, the ninth embodiment realizes a novel multi-level signal transmission system by combining an OFDM and a C-CDM, which was not obtained by the OFDM alone.

An OFDM is certainly strong against multipath such as TV ghost because the guard time Tg can absorb an interference signal of multipath. Accordingly, the OFDM is applicable to the digital TV broadcasting for automatic vehicle TV receivers. Meanwhile, no OFDM signal is received when the C/N rate is less than a predetermined value because its signal transmission pattern is non not of a multi-level type.

However the present invention can solve this disadvantage by combining the OFDM with the C-CDM, thus realizing a gradational degradation depending on the C/N rate in a video signal reception without being disturbed by multipath.

When a TV signal is received in a compartment of vehicle, not only the reception is disturbed by multipath but the C/N rate is deteriorated. Therefore, the broadcast service area of a TV broadcast station will not be expanded as expected if the countermeasure is only for multipath.

On the other hand, a reception of TV signal of at least LDTV grade will be ensured by the combination with the multi-level transmission C-CDM even if the C/N rate is fairly deteriorated. As a picture plane size of an automotive vehicle TV is normally less than 10 inches, a TV signal of an LDTV grade will provide a satisfactory picture quality. Thus, the LDTV grade service area of automotive vehicle TV will be largely expanded. If an OFDM is used in an entire frequency band of HDTV signal, present semiconductor technologies cannot prevent circuitry scale from increasing so far.

Now, an OFDM method of transmitting only D_1-1 of low frequency band TV signal will be explained below. As shown in a block diagram in FIG. 138, a medium frequency band component D_1-2 and a high frequency band component D₂ of an HDTV signal are multiplexed in C-CDM modulator 4a, and then transmitted at a frequency band A through an FDM 40d.

On the other hand, a signal received by a receiver 43 is first of all frequency separated by an FDM 40e and, then, demodulated by a C-CDM demodulator 4b of the present invention. Thereafter, thus C-CDM demodulated signal is reproduced into medium and high frequency components of HDTV in the same way as in FIG. 123. An operation of a video decoder 402 is identical to that of embodiments 1, 2, and 3 and will no more be not be further explained.

Meanwhile, the D_1-1 signal, a low frequency band signal of MPEG 1 grade of HDTV, is converted by a serial to parallel converter 791 into a parallel signal and fed to an OFDM modulator 852c, which executes QPSK or 16 QAM modulation. Subsequently, the D_1-1 signal is converted by an inverse FFT 40 into a time-base signal and transmitted at a frequency band B through a FDM 40d.

On the other hand, a signal received by the receiver 43 is frequency separated in the FDM 40e and, then, converted into a number of frequency-base signals in an FFT 40a of an OFDM modulator demodulator 852d. Thereafter, frequency-base signals are demodulated in respective demodulators 4a, 4b, - - - and are fed into a parallel to serial converters 882a, wherein a D_i D_1-1 signal is demodulated. Thus, a D_1-1 signal of LDTV grad is outputted from the receiver 43.

In this manner, only an LDTV signal is OFDM modulated in the multi-level signal transmission. The method of FIG. 138 makes it possible to provide a complicated OFDM circuit only for an LDTV signal. A bit rate of LDTV signal is 1/20 of that of an HDTV. Therefore, the circuit scale of the OFDM will be reduced to 1/20, which results in an outstanding reduction of overall circuit scale.

An OFDM signal transmission system is strong against multipath and will soon be applied to a moving station, such as a portable TV, an automotive vehicle TV, or a digital music broadcast receiver, which is exposed under strong and variable multipath obstruction. For such usages a small picture size of less than 10 inches, 4 to 8 inches, is the mainstream. It will be thus guessed that the OFDM modulation of a high resolution TV signal such as HDTV or EDTV will bring less effect. In other words, the reception of a TV signal of LDTV grade would be sufficient for an automotive vehicle TV.

On the contrary, multipath is constant at a fixed station such as a home TV. Therefore, a countermeasure against multipath is relatively easy. Less effect will be brought to such a fixed station by OFDM unless it is in a ghost area. Using OFDM for medium and high frequency band components of HDTV is not advantageous in view of present circuit scale of OFDM which is still large.

Accordingly, the method of the present invention, in which OFDM is used only for a low frequency band TV signal as shown in FIG. 138, can widely reduce the circuit scale of the OFDM to less than 1/10 without losing inherent OFDM effect capable of largely reducing multiple obstruction of LDTV when received at a mobile station such as an automotive vehicle.

Although the OFDM modulation of FIG. 138 is performed only for D_1-1 signal, it is also possible to modulate both D_1-1 and D_1-1 D_1-2 by OFDM. In such a case, a C-CDM two-level signal transmission is used for transmission of D_1-1 and D_1-2. Thus, a multi-level broadcasting being strong against multipath will be realized for a vehicle such as an automotive vehicle. Even in a vehicle, the gradational graduation will be realized in such a manner that LDTV and SDTV signals are received with picture qualities depending on receiving signal level or antenna sensitivity.

The multi-level signal transmission according to the present invention is feasible in this manner and produces various effects as previously described. Furthermore, if the multi-level signal transmission of the present invention is incorporated with an OFDM, it will be possible to provide a system strong against multipath and to alter the transmission grade in accordance with receivable signal level change.

FIG. 126(a) shows another method of realizing the multi-level signal transmission system, wherein the subchannels 794a-794c of the OFDM are assigned to a first layer 801a and the subchannels 794d-794f are assigned to a second layer 801b. There is provided a frequency guard zone 802a of f_g between these two, first and second, layers. FIG. 126(b) shows an electric power difference 820b of Pg which is provided to differentiate the transmission power of the first and second layers 801a and 801b.

Utilization of this differentiation makes it possible to increase electric power of the first layer 801a in the range not obstructing the analogue TV broadcast service as shown in FIG. 108(d) previously described. In this case, a threshold value of the C/N ratio capable of receiving the first layer 801a becomes lower than that for the second layer 801b as shown in FIG. 108(e). Accordingly, the first layer 801a can be received even in a low signal-level area or in a large-noise area. Thus, a two-layer signal transmission is realized as shown in FIG. 147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) in this specification. If this PW-OFDM system is combined with the C-CDM system previously explained, three layers will be realized as shown in FIG. 108(e) and, accordingly, the signal receivable area will be correspondingly expanded.

FIG. 144 shows a specific circuit, wherein the first layer data passing through the first data stream circuit 791a is modulated into the carriers f₁-f₃ by the modulators 4a-4c having large amplitude and, then, are OFDM modulated in the inverse FFT 40. On the contrary, the second layer data passing through the second data stream circuit 791b is modulated into the carriers f₆-f₈ by the modulators 4d-4f having ordinary amplitude and, then, are OFDM modulated in the inverse FFT 40. Then, these OFDM modulated signals are transmitted from the transmit circuit 5.

A signal received by the receiver 43 is separated into several signals having carriers of f₁-f_n through the FFT 40a. The carriers f₁-f₃ are demodulated by the demodulators 45a-45c to reproduce the first data stream D₁, i.e. the first layer 801a. On the other hand, the carriers f₆-f₈ are demodulated by the demodulators 45d-45f to reproduce the second data stream D₂, i.e. the second layer 801b.

The first layer 801a has so large electric power that it can be received even in a weak-signal area. In this manner, the PW-OFDM system realizes the two-layer multi-level signal transmission. If this OW-OFDM is combined with the C-CDM, it will become possible to provide 3-4 layers. As the circuit of FIG. 144 is identical with the circuit of FIG. 123 in the remaining operations and, therefore, will no more not be further explained.

Next, a method of realizing a multi-level signal transmission in Time-Weighted-OFDM (i.e. TW-OFDM) in accordance with the present invention will be explained. Although the OFDM system is accompanied with the guard time zone t_S as previously described, adverse affection of ghost will be eliminated if the delay time t_M of the ghost, i.e. multipath, signal satisfies the requirement of t_M<t_g. The delay time t_M will be relatively small, for example in the range of several μs, in a fixed station such as a TV receiver used for home use. Furthermore, as its value is constant, cancellation of ghost will be relatively easily done. On the contrary, reflected wave will increase in case of a mobile station such as a vehicle TV receiver. Therefore, the delay time t_M becomes relatively large, for example in the range of several tens μs. Furthermore, the magnitude of t_M varies in response to the running movement of the vehicle. Thus, cancellation of ghost tends to be difficult. Hence, the multi-level signal transmission is key or essential for such a mobile station TV receiver in order to eliminate adverse affection of multipath.

The multi-level signal transmission in accordance with the present invention will be explained below. A symbol contained in the subchannel layer A can be intensified against the ghost by setting a guard time t_ga of the layer A to be larger than a guard time t_gb of the layer B as shown in FIG. 146. In this manner, the multi-layer signal transmission can be realized against multipath by use of weighting of guard time. This system is referred to as Guard-Time-Weighted-OFDM (i.e. QTW-OFDM).

If the symbol number of the symbol time Ts is not different in the layer A and in the layer B, a symbol time t_sa of the layer A is set to be larger than a symbol time t_sb of the layer B. With this differentiation, a carrier with Δfa of the carrier A becomes larger smaller than a carrier with Δfb of the carrier B. (Δfa> ≦Δfb) Therefore, the error rate becomes a lower in the demodulation of the symbol of the layer A compared with the demodulation of the symbol of the layer B. Thus, the differentiation of the layers A and B in the weighting of the symbol time Ts can realize a two-layer signal transmission against multipath. This system is referred to as Carrier-Spacing-Weighted-OFDM (i.e. CSW-OFDM).

By realizing the two-layer signal transmission based on the GTW-OFDM, wherein a low-resolution TV signal is transmitted by the layer A and a high-frequency component is transmitted by the layer B, the vehicle TV receiver can stably receive the low-resolution TV signal regardless of tough ghost. Furthermore, the multi-level signal transmission with respect to the C/N ratio can be realized by differentiating the symbol time t_s based on the CSW-OFDM between the layers A and B. If this CSW-OFDM is combined with the GTW-OFDM, the signal reception in the vehicle TV receiver can be further stabilized. High resolution is not normally required to the vehicle TV or the portable TV.

As the time ratio of the symbol time including a low-resolution TV signal is small, an overall transmission efficiency will not decrease so much even if the guard time is enlarged. Accordingly, using the GTW-OFDM of the present invention for suppressing multipath by laying emphasis on the low-resolution TV signal will realize the multi-layer type TV broadcast service wherein the mobile station such as the portable or vehicle TV receiver can be compatible with the stationary station such as the home TV without substantially lowering the transmission efficiency. If combined with the CSW-OFDM or the C-CDM as described previously, the multi-layer to the C/N ratio can be also realized. Thus, the signal reception in the mobile station will be further stabilized.

An affection of the multipath will be explained in more detail. In case of multipaths 810a, 810b, 810c, and 810d having shorter delay time as shown in FIG. 145(a), the signals of both the first and second layers can be received and therefore the HDTV signal can be demodulated. On the contrary, in case of multipaths 811a, 811b, 811c, and 811d having longer delay time as shown in FIG. 145(b), the B signal of the second layer cannot be received since its guard time t_gb is not sufficiently long. However, the A signal of the first layer can be received without being bothered by the multipath since its guard time t_ga is sufficiently long. As described above, the B signal includes the high-frequency component of TV signal. The A signal includes the low-frequency component of TV signal. Accordingly, the vehicle TV can reproduce the LDTV signal. Furthermore, as the symbol time Tsa is set larger than symbol time Tsb, the first layer is strong against deterioration of C/N ratio.

Such a discrimination of the guard time and the symbol time is effective to realize two-dimensional multi-layer signal transmission of the OFDM in a simple manner. If the discrimination of guard time is combined with the C-CDM in the circuit shown in FIG. 123, the multi-layer signal transmission effective against both multipath and deterioration of C/N ratio will be realized.

Next, a specific example will be described below.

The smaller the D/U ratio of the receiving signal becomes, the larger the multipath delay time T_M becomes. Because, the reflected wave increases compared with the direct wave. For example, as shown in FIG. 148, if the D/U ratio is smaller than 30 dB, the delay time T_M exceeds 30 μs because of increase of the reflected wave. Therefore, as can be understood from FIG. 148, it will become possible to receive the signal even in the worst condition if the Tg is set to be larger than 50 μs.

Accordingly, as shown in detail in FIGS. 149(a) and 149(b), three groups of first 801a, second 801b, and third 801c layers are assigned in a 2 ms period of 1 sec TV signal. The guard times 797a, 797b, and 797c, i.e. Tga, Tgb, and Tgc, of these three groups are weighted to be, for example, 50 μs, 5 μs, and 1 μs, respectively, as shown in FIG. 149(c). Thus, three-layer signal transmission effective to the multipath will be realized as shown in FIG. 150, wherein three layers 801a, 801b, and 801c are provided.

If the GTW-OFDM is applied to all the picture quality, it is doubtless that the transmission efficiency will decrease. However, if the GTW-OFDM is only applied to the LDTV signal including less information for the purpose of suppression of multipath, it is expected that an overall transmission efficiency will not be worsened so much. Especially, as the first layer 801a has a long guard time Tg of 50 μs larger than 30 μs, it will be received even by the vehicle TV receiver. The circuit shown in FIG. 127 will be suitable for this purpose. Especially, the requirement to the quality of vehicle TV is LDTV grade. Therefore, its transmission capacity will be approximately 1 Mbps of MPEG 1 class. If the symbol time 796a, i.e. Tsa, is set to be 200 μs with respect to the 2 ms period as shown in FIG. 149, the transmission capacity becomes 2 Mbps. Even if the symbol rate is lowered less than half, an approximately 1 Mbps capacity can be kept. Therefore, it is possible to ensure picture quality of LDTV grade. Although the transmission efficiency is slightly decreased, the error rate can be effectively lowered by the CSW-OFDM in accordance with the present invention. If the C-CDM of the present invention is combined with the GTW-OFDM, deterioration of the transmission efficiency will be able to be effectively prevented. In FIG. 149, the symbol times 796a, 796b, and 796c of the same symbol number are differentiated to be 200 μs, 150 μs, and 100 μs, respectively. Accordingly, the error rate becomes high in the order of the first, second, and third layers so as to realize the multi-layer signal transmission.

At the same time, the multi-layer signal transmission effective to C/N ratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, a two-dimensional multi-layer signal transmission is realized with respect to the multipath and the C/N ratio as shown in FIG. 151. As described previously, it is possible to combine the CSW-OFDM and the C-CDM of the present invention for preventing the overall transmission efficiency from being lowered. In the first, 1-2, and 1-3 layers, 801a, 851a, and 851az, the LDTV grade signal can be stably received by, for example, the vehicle TV receiver subjected to the large multipath T_M and low C/N ratio. In the second and 2-3 layers 801b and 851b, the standard-resolution SDTV grade signal can be received by the fixed or stationary station located, for example, in the fringe of the service area which is generally subjected to the lower C/N ratio and ghost. In the third layer 801c which occupies more than half of the service area, the HDTV grade signal can be received since the C/N ratio is high and the ghost is less because of large direct wave. In this manner, a two-dimensional multi-layer broadcast service effective to both the C/N ratio and the multipath can be realized by the combination of the GTW-OFDM and the C-CDM or the combination of the GTW-OFDM and the CSW-C-CDM in accordance with the present invention. Thus, the present invention realizes a two-dimensional, matrix type, multi-layer signal transmission system effective to both the C/N ratio and thel the multipath, which has not ever been realized by the prior art technologies.

A timing chart of a three level (HDTV, SDTV, LDTV) television signal in a two-dimensional multilevel broadcast of three C/N levels and three multipath levels is shown in FIG. 152. As shown in the figure, the LDTV signal is positioned in slot 796a1 of the first level of level layer A, the level with the greatest resistance to multipath interference; the SDT synchronization signal, address signal, and other important high priority signals are positioned in slot 796a2, which has the next greatest resistance to multipath interference, and slot 796b1, which has strong resistance to C/N deterioration. The SDTV common signal, i.e., low priority signals, and HDTV high priority signals are positioned in levels 2 and 3 of level B. SDTV, EDTV, HDTV, and other high frequency component television signals are positioned in levels 1, 2, and 3 of level C.

As the resistance to C/N deterioration and multipath interference increases, the transmission raze rate drops, causing the TV signal resolution to drop, and achieving the three-dimensional graceful degradation effect shown in FIG. 153 and unobtainable with conventional methods. As shown in FIG. 153, the three-dimensional multilevel broadcast structure of the invention is achieved with three parameters: C/N ratio, multipath delay time, and the transmission rate.

The present embodiment has been described using the example of a two-dimensional multilevel broadcast structure obtained by combining GTW-OFDM of the invention with C-CDM of the invention as previously described, or combining GTW-OFDM, CSW-C-CDM, but other two-dimensional multilevel broadcast structures can be obtained by combining GTW-OFDM and power-weighted OFDM, or GTW-OFDM with other C/N ratio multilevel transmission methods.

FIG. 154 is obtained by transmitting the power of carriers 794a, 794c, and 794e with less weighting compared with carriers 794b, 794d, and 794f, achieving a two level power-weighted OFDM. Two levels are obtained by power weighting carriers 795a and 795c, which are perpendicular to carrier 794a, to carriers 795b and 795d. While a total of four levels are obtained, the embodiment having only two levels is shown in FIG. 154. As shown in the figure, because the carrier frequencies are distributed, interference with other analog transmissions on the same frequency band is dispersed, and there is minimal adverse effect.

By using a time positioning varying the time width of guard times 797a, 797b, and 797c for each symbol 796a, 796b, and 796c as shown in FIG. 155, three-level multipath multilevel transmission can be achieved. Using the time positioning shown in FIG. 155, the A-, B-, and C -level data is distributed on the time axis. As a result, even if burst noise produced at a specific time occurs, data destruction can be prevented and the TV signal can be stably demodulated by interleaving the data from the different layers. In particular, by interleaving with the A level data distributed, interference from burst noise generated by the ignition systems of the other vehicles can be significantly reduced in mobile TV receivers.

Block diagrams of a specific ECC encoder 744j and a specific ECC decoder 749j 759j are shown in FIG. 160a and FIG. 160b, respectively. FIG. 167 is a block diagram of the deinterleaver 936b. The interleave table 954 processed in the deinterleaver RAM 936a of the deinterleaver 936b is shown in FIG. 168a, and interleaves distance L1 is shown in FIG. 168b.

Burst noise interference can be reduced by interleaving the data in this way. By using a 4-level VSB, 8-level VSB, or 16-level VSB transmission apparatus as described in embodiments 4, 5, and 6, respectively, and shown in the VSB receiver block diagram (FIG. 161) and the VSB transmitter block diagram (FIG. 162), or by using a QAM or PSK transmission apparatus as described in embodiments 1 and 2, respectively, burst noise interference can be reduced, and television reception with very low noise levels can be achieved in ground station broadcasting.

By using 3-level broadcasting by means Of of the method shown in FIG. 155, LDTV grade television reception by mobile receivers, including mobile TV receives in motor vehicles and hand-held portable television sets, can be stabilized because level A has the effect of reducing burst noise interference in addition to multipath interference and C/N ratio deterioration.

The multi-level signal transmission method of the present invention is intended to increase the utilization of frequencies but may be suited for not all the transmission systems since causing some type receivers to be declined in the energy utilization, it is a good idea for use with a satellite communications system for selected subscribers to employ most advanced transmitters and receivers designed for best utilization of applicable frequencies and energy. Such a specific purpose signal transmission system will not be bound by the present invention.

The present invention will be advantageous for use with a satellite or terrestrial broadcast service which is essential to run in the same standards for as long as 50 years. During the service period, the broadcast standards must not be altered but improvements will be provided time to time corresponding to up-to-date technological achievements. Particularly, the energy for signal transmission will surely be increased on any satellite. Each TV station should provide a compatible service for guaranteeing TV program signal reception to any type receivers ranging from today's common ones to future and advanced ones. The signal transmission system of the present invention can provide a compatible broadcast service of both the existing NTSC and HDTV systems and also, ensure a future extension to match mass date data transmission.

The present invention concerns much on the frequency utilization than the energy utilization. The signal receiving sensitivity of each receiver is arranged different differently depending on a signal state level to be retrieved so that the transmitting power of a transmitter needs not be increased largely. Hence, existing satellites which offer a small energy for reception and transmission of a signal can best be used with the system of the present invention. The system is also arranged for performing the same standards corresponding to an increase in the transmission energy in the future and offering the compatibility between old and new type receivers. In addition, the present invention will be more advantageous for use with the satellite broadcast standards.

The multi-level signal transmission method of the present invention is more preferably employed for terrestrial TV broadcast service in which the energy utilization is not crucial, as compared with satellite broadcast service. The results are such that the signal attenuating regions in a service area which are attributed to a conventional digital HDTV broadcast system are considerably reduced in extension and also, the compatibility of an HDTV receiver or display with the existing NTSC system is obtained. Furthermore, the service area is substantially increased so that program suppliers and sponsors can appreciate more viewer viewers. Although the embodiments of the present invention refer to 16 and 32 QAM procedures, other modulation techniques including 64, 128, and 256 QAM will be employed with equal success. Also, multiple PSK, ASK, and FSK techniques will be applicable as described with the embodiments.

A combination of the TDM with the SRQAM of the present invention has been described in the above. However, the SRQAM of the present invention can be combined also with any of the FDM, CDMA and frequency dispersal communications systems.

Claims

1. A digital TV receiver comprising:

a receiving section for receiving a PSK (Phase Shift Key) modulation signal comprising a plurality of signal points disposed on specific phases of a given constellation in a signal space diagram;

a demodulator for demodulating the received signal from said receiving section into a digital signal;

an error correcting section for error correcting a demodulation signal from said demodulator; and

an image expander for expanding an error-corrected signal from said error correcting section to a video signal, thereby outputting a video signal,

wherein said demodulator demodulates the receiving signal as first and second PSK signals, said first PSK signal representing a first data stream to be reproduced, said first PSK signal comprising n-value signal points, said second PSK signal representing both said first data steam and a second data stream to be reproduced, said second PSK signal comprising m-value signal points, where m is an integer larger than n,

wherein the m-value signal points of said second PSK signal are divisible into n groups of signal points which are distinguishable from one another in the signal space, and said demodulator distinguishes said n groups of signal points from one another as the n-value signal points of said first PSK signal by demodulating the received signal as said first PSK signal, and wherein high priority information is demodulated at least in said first data stream.

2. The digital TV receiver in accordance with claim 1, wherein said demodulator demodulates information relating to a low-resolution component of the video signal from said first data stream and demodulates information relating to a high-resolution component of the video signal from said second data stream.

3. The digital TV receiver in accordance with claim 1, comprising means for stopping output of said second data stream when an error rate of the received signal is high.

4. The digital TV receiver in accordance with claim 1, wherein said demodulator comprises first means for demodulating the received signal as a QPSK signal to reproduce the first data stream and second means for demodulating the received signal as a 8PSK signal to reproduce both of the first data stream and the second data stream.

5. A signal TV receiver comprising:

a receiving section for receiving a PSK (Phase Shift Key) modulation signal comprising m-value signal points disposed on specific phases of a given constellation in a signal space diagram and representing a first data stream and a second data stream to be reproduced;

a demodulator for demodulating the said PSK modulation signal from said receiving section into a digital signal;

an error correcting section for error correcting a demodulation signal from said demodulator; and

an image expander for expanding an error-corrected signal from said error correcting section to a video signal,

wherein said demodulator includes means for demodulating said PSK signal comprising m-value signal points as another PSK signal,

said another PSK signal comprising n-value signal points and representing only said first data stream, where n is an integer smaller than m, and wherein

high priority information is demodulated at least in said first data stream.

6. A signal transmission apparatus comprising:

a modulator operable to assign and interleave a data stream of a layer A and a data stream of a layer b to a respective constellation in a signal space to produce a modulated signal of the layer A and a modulated signal of the layer b;

an inverse fast fourier transformer operable to convert the modulated signal of the layer A into a transmission signal on a time axis in the layer A and to convert the modulated signal of the layer b into a transmission signal on a time axis in the layer b, wherein each transmission signal comprises an effective symbol signal and a guard interval signal, and a period of the guard interval signal in the layer A is larger than the period of the guard interval signal in the layer b; and

a transmitter operable to transmit the transmission signals.

7. A signal receiving apparatus for use in receiving a modulation signal in a layer A and a modulation signal in a layer b, wherein the modulation signals each include a guard interval signal, said apparatus comprising:

a fast-fourier transformer operable to convert the modulation signal in the layer A into a converted signal on a frequency axis in the layer A and to convert the modulation signal in the layer b into a converted signal on a frequency axis in the layer b; and

a demodulator operable to de-interleave the converted signals, demodulate the converted signal in the layer A into a data stream of the layer A and to demodulate the converted signal in the layer b into a data stream of the layer b;

wherein a period of the guard interval signal in the layer A is larger than a period of the guard interval signal in the layer b.

8. A signal transmission system comprising:

a signal transmission apparatus including

a modulator operable to assign and interleave a data stream of a layer A and a data stream of a layer b to a respective constellation in a signal space to produce a modulated signal of the layer A and a modulated signal of the layer b,

an inverse fast fourier transformer operable to convert the modulated signal of the layer A into a transmission signal on a time axis in the layer A and to convert the modulated signal of the layer b into a transmission signal on a time axis in the layer b, wherein each transmission signal comprises an effective symbol signal and a guard interval signal, and a period of the guard interval signal in the layer A is larger than a period of the guard interval signal in the layer b, and

a transmitter operable to transmit the transmission signals; and

a signal receiving apparatus including

a receiver operable to receive the transmission signal in the layer A and the transmission signal in the layer b,

a fast-fourier transformer operable to convert the transmission signal in the layer A into a converted signal on a frequency axis in the layer A and to convert the transmission signal in the layer b into a converted signal on a frequency axis in the layer b, and

a demodulator operable to de-interleave the transmission signals, demodulate the converted signal in the layer A into a data stream of the layer A and to demodulate the converted signal in the layer b into a data stream of the layer b.

9. A signal transmission method comprising:

assigning and interleaving a data stream of a layer A and a data stream of a layer b to a respective constellation in a signal space to produce a modulated signal of the layer A and a modulated signal of the layer b;

inverse fast fourier transforming the modulated signal of the layer A into a transmission signal on a time axis in the layer A and inverse fast fourier transforming the modulated signal of the layer b into a transmission signal on a time axis in the layer b, wherein each transmission signal comprises an effective symbol signal and a guard interval signal, and a period of the guard interval signal in the layer A is larger than a period of the guard interval signal in the layer b; and

transmitting the transmission signals.

10. A signal receiving method for use in receiving a modulation signal in a layer A and a modulation signal in a layer b, wherein the modulation signals each include a guard interval signal, and a period of the guard interval signal in the layer A is larger than a period of the guard interval signal in the layer b, said method comprising:

fast-fourier transforming the modulation signal in the layer A into a converted signal on a frequency axis in the layer A and fast-fourier transforming the modulation signal in the layer b into a converted signal on a frequency axis in the layer b; and

de-interleaving the modulation signals, demodulating the converted signal in the layer A into a data stream of the layer A and demodulating the converted signal in the layer b into a data stream of the layer b.