Communication system

Abstract

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into low and high frequency band components which are designated as first and second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

Description

CROSS-REFERENCE TO RELATED APPLICATIONS

transmission32
|A(16)|₂=(A₁+A₂)²+(B₁+B₂)²=4A²_TO+4A²_TO=8A_TO
|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greater amplitude and a four times greater transmitting energy than those needed for the 4 PSK signal. A modified 16 QAM signal according to the present invention will not be demodulated by a common receiver designed for symmetrical, equally distanced signal point QAM. However, it can be demodulated with the second receiver 33 when two threshold values A₁ and A₂ are preset to appropriate values. In FIG. 10, the minimum distance between two signal points in the first segment of the signal point group 91 is A₁ and A₂/2A₁ is established as compared with the distance 2A₁ of 4 PSK. Then, as A₁=A₂, the distance becomes ½. This explains that the signal receiving sensitivity has to be two times greater for the same error rate and four times greater for the same signal level. For having a four times greater value of sensitivity, the radius r₂ of the antenna 32 of the second receiver 33 has to be two times greater than the radius r₁ of the antenna 22 of the first receiver 23 thus satisfying r₂=2r₁. For example, the antenna 32 of the second receiver 33 is 60 cm diameter when the antenna 22 if the first receiver 23 is 30 cm. In this manner, the second data stream representing the high frequency component of an HDTV will be carried on a signal channel and demodulated successfully. As the second receiver 33 intercepts the second data stream or a higher data signal, its owner can enjoy a
Hence, A₂<1.23A₁.

If the distance between any two signal point group segments shown in FIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) and A(16)-A(4) are proportional to

This relationship between between

This relationship is also denoted by the curve 211 in FIG. 16. For example, if the transmitting energy is 6 or 9 times greater than that for 4 PSK transmission at the point 223 or 222, the antenna 32 having a radius of 8× or 6× value respectively can intercept the first, second, and third data streams for demodulation. As the signal point distance of the second data stream is close to ⅔A₂, the relationship between r₁ and r₂ is expressed by:


Therefore, the antenna 32 of the second receiver 33 has to be slightly increased in radius as denoted by the curve 223.

As understood, while the first and second data streams are transmitted through a traditional satellite which provides a small signal transmitting energy, the third data stream can also be transmitted through a future satellite which provides a greater signal transmitting energy without interrupting the action of the first and second receivers 23 or 33 or with no need of modification of the same and thus, both the compatibility and the advancement is ensured.

The signal receiving action of the second receiver 33 will first be described. As compared with the first receiver 23 arranged for interception with a small radius r₁ antenna and demodulation of the 4 PSK modulated signal of the digital transmitter 51 or the first data stream of the signal of the transmitter 1, the second receiver 33 is adopted for perfectly demodulating the 16 signal state two-bit data, shown in FIG. 10, or second data stream of the 16 QAM signal from the transmitter 1. In total, four-bit data including also the first data stream can be demodulated. The ratio between A₁ and A₂ is however different in the two transmitters. The two different data are loaded to a demodulation controller 231 of the second receiver 33, shown in FIG. 21, which in turn supplies their respective threshold values to the demodulating circuit for AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 is similar in basic construction to that of the first receiver 23 shown in FIG. 19. The difference is that the radius r₂ of the antenna 32 is greater than r₁ of the antenna 22. This allows the second receiver 33 to identify a signal component involving a smaller signal point distance. The demodulator 35 of the second receiver 33 also contains first and second data stream reproducing units 232 and 233 in addition to the demodulation controller 231. There is provided a first discrimination/demodulation circuit 136 for AM demodulation of modified 16 QAM signals. As understood, each carrier is a four-bit signal having two, positive and negative, threshold values about the zero level.

The various data for demodulation including A₁ and A₂ or TH₁₆, and the value m for multiple-bit modulation are also transmitted from the transmitter 1 as carried in the first data stream. The demodulation controller 231 may be arranged for recovering such demodulation data through statistical process of the received signal.

A way of determining the shift factor A₁/A₂ will be described with reference to FIG. 26. A change of the shift factor A₁/A₂ causes a change of the threshold value. Increase of a difference of a value of A₁/A₂ set at the receiver side from a value of A₁/A₂ set at the transmitter side will increase the error rate. Referring to FIG. 26, the demodulated signal from the second data stream reproducing unit 233 may be fed back to the demodulation controller 231 to change the shift factor A₁/A₂ in a direction to increase the error rate. By this arrangement, the third receiver 43 may not demodulate the shift factor A₁/A₂, so that the circuit construction can be simplified. Further, the transmitter may not transmit the shift factor A₁/A₂, so that the transmission capacity can be increased. This technique can be applied also to the second receiver 33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for the C-CDM signal points, wherein signal points are added by shifting in the polar coordinate direction (r, θ). The previously described C-CDM is characterized in that the signal points are shifted in the rectangular coordinate direction, i.e. XY direction; therefore it is referred to as rectangular coordinate system C-CDM. Meanwhile, this C-CDM characterized by the shifting of signal points in the polar coordinate direction, i.e. r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein four signal points are added by shifting each of 4 QPSK signals in the radius r direction of the polar coordinate system. In this manner, the APSK of polar coordinate system C-CDM having 8 signal points is obtained from the QPSK as shown in FIG. 25(a). As the pole is shifted in the polar coordinate system to add signal points in this APSK, it is referred to as shifted pole-APSK, i.e. SP-APSK in the abbreviated form. In this case, coordinate values of the newly added four QPSK signals 85 are specified by using a shift factor S₁ as shown in FIG. 139. Namely, 8PS-APSK signal points includes ordinary QPSK signal points 83 (r₀, θ₀) and a signal point ((S₁+1)(r₀, θ₀) obtained by shifting the signal point 83 in the radius r direction by an amount of S₁r₀. Thus, a 1-bit subchannel 2 is obtained in addition to a 2-bit subchannel 1 identical with the QPSK.

Furthermore, as shown in the constellation diagram of FIG. 140, new eight signal points, represented by coordinates (r₀+S₂r₀, θ₀) and (r₀+S₁r₀+S₂r₀, θ₀), can be added by shifting the eight signal points (r₀, θ₀) and (r₀+S₁r₀, θ₀) in the radius r direction. As this allows two kinds of allocations, a 1-bit subchannel is obtained and is referred to as 16PS-APSK which provides the 2-bit subchannel 1, 1-bit subchannel 2, and 1-bit subchannel 3. As the 16-PS-APSK disposes the signal points on the lines of θ=¼(2n+1)π, it allows the ordinary QPSK receiver explained with reference to FIG. 19 to reproduce the carrier wave to demodulate the first 2-bit subchannel although the second subchannel cannot be demodulated. As described above, the C-CDM method of shifting the signal points in the polar coordinate direction is useful in expanding the capacity of information data transmission while assuring compatibility to the PSK, especially to the QPSK receiver, a main receiver for the present satellite broadcast service. Therefore, without losing the first generation viewers of the satellite broadcast service based on the PSK, the broadcast service will advance to a second generation stage wherein the APSK will be used to increase transmittable information amount by use of the multi-level modulation while maintaining compatibility.

In FIG. 25(b), the signal points are allocated on the lines of θ=π/8. With this arrangement, the 16 PSK signal points are reduced or limited to 12 signal points, i.e. 3 signal points in each quadrant. With this limitation, these three signal points in each quadrant are roughly regarded as one signal point for 4 QPSK signals. Therefore, this enables the QPSK receiver to reproduce the first subchannel in the same manner as in the previous embodiment.

More specifically, the signal points are disposed on the lines of θ=π/4, θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offset by an amount ±θ in the angular direction of the polar coordinate system from the QPSK signals disposed on the lines of θ=π/4. Since all the signals are in the range of θ=π/4±π/8, they can be regarded as one of the QPSK signal points on the line of θ=π/4. Although the error rate is lowered a little bit in this case, the QPSK receiver 23 shown in FIG. 19 can discriminate these points as four signal points angularly allocated. Thus, 2-bit data can be reproduced. In case of the angular shift C-CDM, if signal points are disposed on the lines of π/n, the carrier wave reproduction circuit can reproduce the carrier wave by the use of an n-multiplier circuit in the same manner as in other embodiments. If the signal points are not disposed on the lines of π/n, the carrier wave can be reproduced by transmitting several pieces of carrier information within a predetermined period in the same manner as in other
This calculation is equivalent to that of TH₁₆ but its resultant distance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A₃, the distance between the first 181 and the second sub segment 182 is expressed by (A₂−2A₃). Then, the average distance is (A₂−2A₃)/(A₁+A₂) which is designated as d₆₄. when d₆₄ is smaller than T₂ which represents the signal point discrimination capability of the second receiver 33, any two signal points in the segment will hardly be distinguished from each other. This judgement is executed at Step 313. If d₆₄ is out of a permissive range, the procedure moves back to Step 313 for 4 PSK mode demodulation. If d₆₄ is within the range, the procedure advances to Step 305 for allowing the demodulation of 16 QAM at Step 307. If it is determined at Step 308 that the error rate is too high, the procedure goes back to Step 313 for 4 PSK mode demodulation.

When the transmitter 1 supplied a modified 8 QAM signal such as shown in FIG. 25(a) in which all the signal points are at angles of cos (2πf+·π/4), the carrier waves of the signal are lengthened to the same phase and will thus be reproduced with much ease. At the time, two-bit data of the first data stream are demodulated by the 4-PSK receiver while one-bit data of the second data stream is demodulated by the second receiver 33 and the total of three-bit data can be reproduced.

The third receiver 43 will be described in more detail. FIG. 26 shows a block diagram of the third receiver 43 similar to that of the second receiver 33 in FIG. 21. The difference is that a third data stream reproducing unit 234 is added and also, the discrimination/demodulation circuit has a capability of identifying eight-bit data. The antenna 42 of the third receiver 43 has a radius r₃ greater than r₂ thus allowing smaller distance state signals, e.g. 32- or 64-state QAM signals, to be demodulated. For demodulation of the 64 QAM signal, the first discrimination/ reproduction circuit 136 has to identify 8 digital levels of the detected signal in which seven different threshold levels are involved. As one of the threshold values is zero, three are contained in the first quadrant.

FIG. 27 shows a space diagram of the signal in which the first quadrant contains three different threshold values.

As shown in FIG. 27, when the three normalized threshold values are TH1₆₄, TH2₆₄, and TH3₆₄ they are expressed by:
TH1₆₄=(A₁+A₃/2)/(A₁+A₂)
TH2₆₄=(A₁+A₂/2)/(A₁+A₂) and
TH3₆₄=(A₁+A₂−A₃/2)/(A₁+A₂)

Through AM demodulation of a phase detected signal using the three threshold values, the third data stream can be reproduced like the first and second data stream explained with FIG. 21. The third data stream contains e.g. four signal points 201, 202, 203, and 204 at the first sub segment 181 shown in FIG. 23 which represent 4 values of two-bit pattern. Hence, six digits or modified 64 QAM signals can be demodulated.

The demodulation controller 231 detects the value m, A₁, A₂, and A₃ from the demodulation data contained in the first data stream demodulated by the first data stream reproducing unit 232 and calculates the three threshold values TH1₆₄, TH2₆₄, and TH3₆₄ which are then fed to the first 136 and the second discrimination/demodulation circuit 137 so that the modified 64 QAM signal is demodulated with certainty. Also, if the demodulation data have been scrambled, the modified 64 QAM signal can be demodulated only with a specific or subscriber receiver. FIG. 28 is a flowchart showing the action of the demodulation controller 231 for modified 64 QAM signals. The difference from the flowchart for demodulation of 16 QAM shown in FIG. 24 will be explained. The procedure moves from Step 304 to Step 320 where it is determined whether or not m=32 or not. If m=32, demodulation of 32 QAM signals is executed at Step 322. If not, the procedure moves to Step 321 where it is determined whether or not m=64. If yes, A₃ is examined at Step 323. If A₃ is smaller than a predetermined value, the procedure moves to Step 305 and the same sequence as of FIG. 24 is implemented. If it is judged at Step 323 that A₃ is not smaller than the predetermined value, the procedure goes to Step 324 where the threshold values are calculated. At Step 325, the calculated threshold values are fed to the first and second discrimination/demodulation circuits and at Step 326, the demodulation of the modified 64 QAM signal is carried out. Then, the first, second, and third data streams are reproduced at Step 327. At Step 328, the error rate is examined. If the error rate is high, the procedure moves to Step 305 where the 16 QAM demodulation is repeated and if low, the demodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of a satisfactory demodulating procedure will now be described. The scope of the present invention includes reproduction of the first data stream of a modified 16 or 64 QAM signal using a 4 PSK receiver. However, a common 4 PSK receiver rarely reconstructs carrier waves, thus failing to perform a correct demodulation. For compensation, some arrangements are necessary at both the transmitter and receiver sides.

Two techniques for compensation are provided according to the present invention. A first technique relates to transmission of signal points aligned at angles of (2n−1)π/4 at intervals of a given time. A second technique offers transmission of signal points arranged at intervals of an angle of nπ/8.

According to the first technique, the eight signal points including 83 and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown in FIG. 38. In action, at least one of the eight signal points is transmitted during sync time slot periods 452, 453, 454, and 455 arranged at equal intervals of time in a time slot gap 451 shown in the time chart of FIG. 38. Any desired signal points are transmitted during the other time slots. The transmitter 1 is also arranged to assign a data for the time slot interval to the sync timing data region 499 of a sync data block, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detail referring to FIG. 41. The time slot group 451 containing the sync time slots 452, 453, 454, and 455 represents a unit data stream or block 491 carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of a given time determined by the time slot interval or sync timing data. Hence, when the arrangement of the sync time slots is detected, reproduction of carrier waves will be executed slot by slot through extracting the sync timing data from their respective time slots. Such a sync timing data S is contained in a sync block 493 at the front end of a data frame 492, which consists of a number of sync time slots denoted by the hatching in FIG. 41. Accordingly, the data to be extracted for carrier wave reproduction are increased, thus allowing the 4 PSK receiver to reproduce desired carrier waves at higher accuracy and efficiency.

The sync block 493 comprises sync data regions 496, 497, and 498, —containing sync data S1, S2, and S3, —respectively which include unique words and demodulation data. The phase sync signal assignment region 499 is at the end of the sync block 493, which holds a data of I_T including information about interval arrangement and assignment of the sync time slots.

The signal point data in the phase sync time slot has a particular phase and can thus be reproduced by the 4 PSK receiver. Accordingly, I_T in the phase sync signal assignment region 499 can be retrieved without error thus ensuring the reproduction of carrier waves being

The other signal points 84a and 86a are also shifted to two points 84 and 86 respectively.

If the error rate of the first data stream is Pe₁, it is obtained from:

$\begin{array}{c}{\mathrm{Pe}}_{1\text{-}16}=\frac{1}{4}\mathrm{erfc}\left(\frac{n\delta }{\sqrt{2\sigma }}\right)+\frac{1}{4}\mathrm{erfc}\left(\frac{3\delta }{\sqrt{2\sigma }}\right)\\ =\frac{1}{8}\mathrm{erfc}\left(\frac{n\sqrt{p}}{\sqrt{9+n^2}}\right)\end{array}$
Also, the error rate Pe₂ of the second data stream is obtained from:

$\begin{array}{c}{\mathrm{Pe}}_{2\text{-}16}=\frac{1}{2}\mathrm{erfc}\left(\frac{\frac{3-n}{2}\delta }{\sqrt{2\sigma }}\right)\\ =\frac{1}{4}\mathrm{erfc}\left(\frac{\frac{3-n}{2}\delta }{2\sqrt{9+n^2}}\sqrt{p}\right)\end{array}$

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is a vector diagram of a 36 SRQAM signal in which the distance between any two 36 QAM signal points is 2δ.

The signal point 83a of 36 QAM is spaced δ from each axis of the coordinate. It is now assumed that n is a shift value of the 16 SRQAM. In 36 SRQAM, the signal point 83a is shifted to a signal point 83 where the distance from each axis is nδ. Similarly, the nine 36 QAM signal points in the first quadrant are shifted to points 83, 84, 85, 86, 97, 98, 99, 100, 101 respectively. If a signal point group 90 comprising the nine signal points is regarded as a single signal point, the error rate Pe₁ in reproduction of only the first data stream D₁ with a modified 4 PSK receiver and the error rate Pe₂ in reproduction of the second data stream D₂ after discriminating the nine signal points of the group 90 from each other, are obtained respectively from:

$\begin{array}{c}{\mathrm{Pe}}_{1\text{-}32}=\frac{1}{6}\mathrm{erfc}\left(\frac{n\delta }{\sqrt{2\sigma }}\right)\\ =\frac{1}{6}\mathrm{erfc}\left(\sqrt{\frac{6p}{5}}\times \frac{n}{\sqrt{n^2+2n+25}}\right)\end{array}$ $\begin{array}{c}{\mathrm{Pe}}_{2\text{-}32}=\frac{2}{3}\mathrm{erfc}\left(\frac{5-n}{4\sqrt{22}}\frac{n}{p}\right)\\ =\frac{2}{3}\mathrm{erfc}\left(\sqrt{\frac{3p}{40}}\times \frac{5-n}{\sqrt{n^2+2n+25}}\right)\end{array}$

FIG. 101 shows the relationship between error rate Pe and C/N rate in transmission in which the curve 900 represents a conventional or not modified 32 QAM signal. The straight line 905 represents a signal having 10^−1.5 of the error rate. The curve 901a represents a D₁ level 32 SRQAM signal of the present invention at the shift rate n of 1.5. As shown, the C/N rate of the 32 SRQAM signal is 5 dB lower at the error rate of 10^−1.5 than that of the conventional 32 QAM. This means that the present invention allows a D₁ signal to be reproduced at a given error rate when its C/N rate is relatively low.

The curve 902a represents a D₂ level SRQAM signal at n=1.5 which can be reproduced at the error rate of 10^−1.5 only when its C/N rate is 2.5 dB higher than that of the conventional 32 QAM of the curve 900. Also, the curves 901b and 902b represent D₁ and D₂ SRQAM signals at n=2.0 respectively. The
Hence, if the 32 SRQAM signal is selected, the shift n is determined by:
1<n<5
Also, if the 16 SRQAM signal is employed, n is determined by:
1<n<3

In the SRQAM mode signal terrestrial broadcast service in which the first and second data levels are created by shifting corresponding signal points as shown in FIGS. 99 and 100, the advantage of the present invention will be given when the shift n in a 16, 32, or 64 SRQAM signal is more than 1.0.

In the above embodiments, the low and high frequency band components of a video signal are transmitted as the first and second data streams. However, the transmitted signal may be an audio signal. In this case, low frequency or low resolution components of an audio signal may be transmitted as the first data stream, and high frequency or high resolution components of the audio signal may be transmitted as the second data stream. Accordingly, it is possible to receive high C/N portion in high sound quality, and low C/N portion in low sound quality. This can be utilized in PCM broadcast, radio, portable telephone and the like. In this case, the broadcasting area or communication distance can be expanded as compared with the conventional systems.

Furthermore, the third embodiment can incorporate a time division multiplexing (TDM) system as shown in FIG. 133. Utilization of the TDM makes it possible to increase the number of subchannels. An ECC encoder 743a and ECC encoder 743b, provided in two subchannels, differentiate ECC code gains so as to make a difference between thresholds of these two subchannels, whereby an increase in the number of channels of the multi-level signal transmission can be realized. In this case, it is also possible to provide the ECC encoder, such as two Trellis encoders 743a and 743b for VSB-ASK signals of 4 VSB, 8 VSB, 16 VSB as shown in FIG. 137 and differentiate their code gains. The explanation of this block diagram is substantially identical to that of later described block diagram of FIG. 131 which shows the sixth embodiment of the present invention and, therefore, will not be described here.

FIG. 131 is a block diagram of the magnetic recording and reproducing apparatus, and FIG. 137 is a block diagram of the transmission apparatus.

The up converter of the transmitter and the down converter of the receiver of the transmission apparatus can be substituted for the magnetic head recording signal amplifier circuit and the magnetic head reproducing signal amplifier circuit of the magnetic recording and reproducing apparatus, respectively, and there respective components are therefore identically constructed. The configuration and operation of the modulator and demodulator of the magnetic recording and reproducing apparatus are therefore also identical to those of the transmission apparatus. Similarly, the recording/reproducing/transmission system shown in FIG. 84 is identical in construction to the transmission system shown in FIG. 156. To further simplify the system, the configuration shown in the block diagram of FIG. 157 can be used, or for even greater simplification the block diagram of FIG. 158 can be used.

In a simulation of FIG. 106, there is provided 5 dB difference of a coding gain between 1-1 subchannel D_1-1 and 1-2 subchannel D_1-2.

An SRQAM is the system applying a C-CDM (Constellation-Code Division Multiplex) of the present invention to a rectangle-QAM. A C-CDM, which is a multiplexing method independent of TDM or FDM, can obtain subchannels by dividing a constellation-code corresponding to a code. An increase of the number of codes will bring an expansion of transmission capacity, which is not attained by TDM or FDM alone, while maintaining almost perfect compatibility with conventional communication apparatus. Thus C-CDM can bring excellent effects.

Although above embodiment combines the C-CDM and the TDM, it is also possible to combine the C-CDM with the FDM (Frequency Division Multiplex) to obtain similar modulation effect of threshold values. Such a system can be used for a TV broadcasting, and FIGS. 108(a)-108(e) show area
Hence, the multi-level signal transmission system of the embodiment is based on L₀>L. The embodiment is however not limited to L₀>L, and L₀=L will be employed temporarily or permanently depending on the requirements of design, condition, and setting. In the case of VSB, the constellation shown in FIGS. 68(a) and (b) are taken.

The two signal point 8 55cbea an OFDM transmitter/receiver, and FIG. 124 is a diagram showing a principle of an OFDM action. An OFDM is one of FDM and has a better efficiency in frequency utilization as compared with a general FDM, because an OFDM sets adjacent two carriers to be quadrate with each other. Furthermore, OFDM can bear multipath obstructions such as ghosts and, therefore, may be applied in the future to the digital music broadcasting or digital TV broadcasting.

As shown in the principle diagram of FIG. 124, OFDM converts an input signal by a serial to parallel converter 791 into a data being disposed on a frequency axis 793 at intervals of 1/ts, so as to produce subchannels 794a^-94e 794a-794e. This signal is inversely FFT converted by a modulator 4 having an inverse 40 into a signal on a time axis 79^˜9 799 to produce a transmission signal 795. This inverse FFT signal is transmitted during an effective symbol period 796 of the time period ts. A guard interval 797 having an amount tg is provided between symbol periods.

A transmitting/receiving action of HDTV signal in accordance with this ninth embodiment will be explained referring to the block diagram of FIG. 123, which shows a hybrid OFDM-CCDM system. An inputted HDTV signal is separated by a video encoder 401 into three-level three-levels, a low frequency band D_1-1, a medium-low frequency band D_1-2, and a high-medium-low frequency band D₂, video signals, and fed into an input section.

In a first data stream input 743, D_1-1 signal is ECC encoded with high code gain and D_1-2 signal is ECC coded with a normal code gain. A TDM 743 performs time division multiplexing of D_1-1 and D_1-2 signals to produce a D₁ signal, which is then fed to a D₁ serial to parallel converter 791d in a modulator 852a. D₁ signal consists of n pieces of parallel data, which are inputted into first inputs of n pieces of C-CDM modulator 4a, 4b,—respectively.

On the other hand, the high frequency band signal D₂ is fed into a second data stream input 744 of the input section 742, in which D₂ signal is ECC (Error Correction Code) encoded in an ECC 744a and then Trellis encoded in a Trellis encoder 744b. Thereafter, the D₂ signal is supplied to a D₂ serial to parallel converter 791b of the modulator 852a and converted into n pieces of parallel data, which are inputted into second inputs of the n pieces of C-CDM modulator 4a, 4b,—respectively.

The C-CDM modulators 4a, 4b, 4c—respectively produces produce 16 SRQAM signal on the basis of D₁ data of the first data stream input and D₂ data of the second data stream input. These n pieces of C-CDM modulator respectively has have a carrier different from each other. As shown in FIG. 124, carriers 79 4a 794a, 794b, 794c,—are arrayed on the frequency axis 793 so that adjacent two carriers are 90°-out-of-phase with each other. Thus C-CDM modulated n pieces of modulated signal are fed into the inverse FFT circuit 40 and mapped from the frequency axis dimension 793 to the time axis dimension 790 799. Thus, time signals 796a, 796b—, having an effective symbol length ts, are produced. There is provided a guard interval zone 797a of Tg seconds between the effective symbol time zones 796a and 796b, in order to reduce multipath obstruction. FIG. 129 is a graph showing a relationship between time axis and signal level. The guard time Tg of the guard interval band 797a is determined by taking account of multipath affection and usage of signal. By setting the guard time Tg longer than the multipath affected time, e.g. TV ghost, modulated signals from the inverse FFT circuit 40 are converted by a parallel to serial converter 4e into one signal and, then, transmitted from a transmitting circuit 5 as an RF signal.

Next, an action of a receiver 43 will be described. A received signal, shown as time-base symbol signal 796e of FIG. 124, is fed into an input section 24 of FIG. 123. Then, the received signal is converted into a digital signal in a demodulator 852b and further changed into Fourier coefficients in a FFT 40a. Thus, the signal is mapped from the time axis 799 to the frequency axis 793a as shown in FIG. 124. That is, the time-base symbol signal is converted into frequency-base carriers 794a, 794b,—. As these carriers are in quadrature relationship with each other, it is possible to separate respective modulated signals. FIG. 125(b) shows thus demodulated 16 SRQAM signal, which is then fed to respective C-CDM demodulators 45a, 45b,—of a C-CDM demodulator 45, in which demodulated 16 SRQAM signal is demodulated into multi-level sub signals D₁, D₂. These sub signals D₁ and D₂ are further demodulated by a D₁ parallel to serial converter 852a and a D₂ parallel to serial converter 852b into original D₁ and D₂ signals.

Since the signal transmission system is of C-CDM multi-level shown in 125(b), both D₁ and D₂ signals will be demodulated under better receiving condition but only D₁ signal will be demodulated under worse, e.g. low C/N rate, receiving condition. Demodulated D₁ signal is demodulated in an output section 757. As D_1-1 signal has higher ECC code gain as compared with the D_1-2 signal, an error signal of the D_1-1 signal is reproduced even under worse receiving condition.

The D_1-1 signal is converted by a 1-1 video decoder 402c into a low frequency band signal and outputted as an LDTV, and the D_1-2 signal is converted by a 1-2 video decoder 402d into a medium frequency band signal and outputted as EDTV.

The D₂ signal is Trellis decoded by a Trellis decoder 75 9b 759b and converted by a second video decoder 402b into a high frequency band signal and outputted as an HDTV signal. Namely, an LDTV signal is outputted in case of the low frequency band signal only. An EDTV signal of a wide NTSC grade is outputted if the medium frequency band signal is added to the low frequency band signal, and an HDTV signal is produced by adding low, medium, and high frequency band signals. As well as the previous embodiment, a TV signal having a picture quality depending on a receiving C/N rate can be received. Thus, the ninth embodiment realizes a novel multi-level signal transmission system by combining an OFDM and a C-CDM, which was not obtained by the OFDM alone.

An OFDM is certainly strong against multipath such as TV ghost because the guard time Tg can absorb an interference signal of multipath. Accordingly, the OFDM is applicable to the digital TV broadcasting for automotive vehicle TV receivers. Meanwhile, no OFDM signal is received when the C/N rate is less than a predetermined value because its signal transmission pattern is non not of a multi-level type.

However the present invention can solve this disadvantage by combining the OFDM with the C-CDM, thus realizing a graditional gradational degradation depending on the C/N rate in a video signal reception without being disturbed by multipath.

When a TV signal is received in a compartment of a vehicle, not only the reception is disturbed by multipath but the C/N rate is deteriorated. Therefore, the broadcast service area of a TV broadcast station will not be expanded as expected if the countermeasure is only for multipath.

On the other hand, a reception of TV signal of at least LDTV grade will be ensured by the combination with the multi-level transmission C-CDM even if the C/N rate is fairly deteriorated. As a picture plane size of an automotive vehicle TV is normally less than 10 inches, a TV signal of an LDTV grade will provide a satisfactory picture quality. Thus, the LDTV grade service area of automotive vehicle TV will be largely expanded. If an OFDM is used in an entire frequency band of HDTV signal, present semiconductor technologies cannot prevent circuitry scale from increasing so far.

Now, an OFDM method of transmitting only D_1-1 of low frequency band TV signal will be explained below. As shown in a block diagram in FIG. 138, a medium frequency band component D_1-2 and a high frequency band component D₂ of an HDTV signal are multiplexed in C-CDM modulator 4a, and then transmitted at a frequency band A through an FDM 40d.

On the other hand, a signal received by a receiver 43 is first of all frequency separated by an FDM 40e and, then, demodulated by a C-CDM demodulator 4b of the present invention. Thereafter, thus C-CDM demodulated signal is reproduced into medium and high frequency components of HDTV in the same way as in FIG. 123. An operation of a video decoder 402 is identical to that of embodiments 1, 2, and 3 and will not be explained in greater detail.

Meanwhile, the D_1-1 signal, a low frequency band signal of MPEG 1 grade of HDTV, is converted by a serial to parallel converter 791 into a parallel signal and fed to an OFDM modulator 852c, which executes QPSK or 16 QAM modulation. Subsequently, the D_1-1 signal is converted by an inverse FFT 40 into a time-base signal and transmitted at a frequency band B through a FDM 40d.

On the other hand, a signal received by the receiver 43 is frequency separated in the FDM 40e and then converted into a number of frequency-base signals in an FFT 40a of an OFDM modulator demodulator 852d. Thereafter, frequency-base signals are demodulated in respective demodulators 4a, 4b,—and are fed into a parallel to serial converter 882a, wherein a D_1-1 signal is demodulated. Thus, a D_1-1 signal of LDTV grade is outputted from the receiver 43.

In this manner, only an LDTV signal is OFDM modulated in the multi-level signal transmission. The method of FIG. 138 makes it possible to provide a complicated OFDM circuit only for an LDTV signal. A bit rate of LDTV signal is 1/20 of that of an HDTV. Therefore, the circuit scale of the OFDM will be reduced to 1/20, which results in an outstanding reduction of overall circuit scale.

An OFDM signal transmission system is strong against multipath and will soon be applied to a moving station, such as a portable TV, an automotive vehicle TV, or a digital music broadcast receiver, which is exposed under strong and variable multipath obstruction. For such usages a small picture size of less than 10 inches, 4 to 8 inches, is the mainstream. It will be thus guessed that the OFDM modulation of a high resolution TV signal such as HDTV or EDTV will bring less effect. In other words, the reception of a TV signal of LDTV grade would be sufficient for an automotive vehicle TV.

On the contrary, multipath is constant at a fixed station such as a home TV. Therefore, a countermeasure against multipath is relatively easy. Less effect will be brought to such a fixed station by OFDM unless it is in a ghost area. Using OFDM for medium and high frequency band components of HDTV is not advantageous in view of present circuit scale of OFDM which is still large.

Accordingly, the method of the present invention, in which OFDM is used only for a low frequency band TV signal as sown shown in FIG. 138, can widely reduce the circuit scale of the OFDM to less than 1/10 without losing inherent OFDM effect capable of largely reducing multiple obstruction of LDTV when received at a mobile station such as an automotive vehicle.

Although the OFDM modulation of FIG. 138 is performed only for D_1-1 signal, it is also possible to modulate both D_1-1 and D_1-1 D_1-2 by OFDM. In such a case, a C-CDM two-level signal transmission is used for transmission of D_1-1 and D_1-2. Thus, a multi-level broadcasting being strong against multipath will be realized for a vehicle such as an automotive vehicle. Even in a vehicle, the gradational graduation will be realized in such a manner that LDTV and SDTV signals are received with picture qualities depending on receiving signal level or antenna sensitivity.

The multi-level signal transmission according to the present invention is feasible in this manner and produces various effects as previously described. Furthermore, if the multi-level signal transmission of the present invention is incorporated with an OFDM, it will become possible to provide a system strong against multipath and to alter data transmission grade in accordance with receivable signal level change.

FIG. 126(a) shows another method of realizing the multi-level signal transmission system, wherein the subchannels 794a-794c of the OFDM are assigned to a first layer 801a and the subchannels 794d-794f are assigned to a second layer 801b. There is provided a frequency guard zone 802a of f_g between these two, first and second, layers. FIG. 126(b) shows an electric power difference 802b of Pg which is provided to differentiate the transmission power of the first and second layers 801a and 801b.

Utilization of this differentiation makes it possible to increase electric power of the first layer 801a in the range not obstructing the analogue TV broadcast service as shown in FIG. 108(d) previously described. In this case, a threshold value of the C/N ratio capable of receiving the first layer 801a becomes lower than that for the second layer 801b as shown in FIG. 108(e). Accordingly, the first layer 801a can be received even in a low signal-level area or in a large-noise area. Thus, a two-layer signal transmission is realized as shown in FIG. 147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) in this specification. If this PW-OFDM system is combined with the C-CDM system previously explained, three layers will be realized as shown in FIG. 108(e) and, accordingly, the signal receivable area will be correspondingly expanded.

FIG. 144 shows a specific circuit, wherein the first layer data passing through the first data stream circuit 791 a is modulated into the carriers f₁-f₃ by the modulators 4a-4c having large amplitude and, then, are OFDM modulated in the inverse FFT 40. On the contrary, the second layer data passing through the second data stream circuit 791b is modulated into the carriers f₆-f₈ by the modulators 4d-4f having ordinary amplitude and, then, are OFDM modulated in the inverse FFT 40. Then, these OFDM modulated signals are transmitted from the transmit circuit 5.

A signal received by the receiver 43 is separated into several signals having carriers of f₁-f_n through the FFT 40a. The carriers f₁-f₃ are demodulated by the demodulators 45a-45c to reproduce the first data stream D₁, i.e. the first layer 801a. On the other hand, the carriers f₆-f₈ are demodulated by the demodulators 45d-45f to reproduce the second data stream D₂, i.e. the second layer 801b.

The first layer 801a has so large electrical power that it can be received even in a weak-signal area. In this manner, the PW-OFDM system realizes the two-layer multi-level signal transmission. If this PW-OFDM is combined with the C-CDM, it will become possible to provide 3-4 layers. As the circuit of FIG. 144 is identical with the circuit of FIG. 123 in the remaining operations and, therefore, will not be explained in greater detail.

Next, a method of realizing a multi-level signal transmission in Time-Weighted-OFDM (i.e. TW-OFDM) in accordance with the present invention will be explained. Although the OFDM System is accompanied with the guard time zone t_g as previously described, adverse affects of ghosts will be eliminated if the delay time t_M of the ghost, i.e. multipath, signal satisfies the requirement of t_M<t_g. The delay time t_M will be relatively small, for example in the range of several microsounds microseconds, in a fixed station such as a TV receiver used for home use. Furthermore, as its value is constant, cancellation of ghosts will be relatively easily done. On the contrary, reflected waves will increase in case of a mobile station such as a vehicle TV receiver. Therefore, the delay time t_M becomes relatively large, for example in the range of several tens microsound microsecond. Furthermore, the magnitude of t_M varies in response to the running movement of the vehicle. Thus, cancellation of ghosts tends to be difficult. Hence, the multi-level signal transmission is key or essential for such a mobile station TV receiver in order to eliminate adverse affection of multipath.

The multi-level signal transmission in accordance with the present invention will be explained below. A symbol contained in the subcannel subchannel layer A can be intensified against the ghosts by setting a guard time t_ga of the layer A to be larger than a guard time t_gb of the layer B as shown in FIG. 146. In this manner, the multi-layer signal transmission can be realized against multipath by use of weighting of guard time. This system is referred to as Guard-Time-Weighted-OFDM (i.e. QTW-OFDM).

If the symbol number of the symbol time Ts is not different in the layer A and in the layer B, a symbol time t_sa of the layer A is set to be larger smaller than a symbol time t_sb of the layer B. With this differentiation, a carrier width Δfa of the carrier A becomes larger than a carrier width Δfb of the carrier B. (Δfa>Δfb) (Δfa<Δfb) Therefore, the error rate becomes lower in the demodulation of the symbol of the layer A compared with the demodulation of the symbol of the layer B. Thus, the differentiation of the layer A and B in the weighting of the symbol time Ts can realize a two-layer signal transmission against multipath. This system is referred to as Carrier-Spacing-Weighted-OFDM (i.e. CSW-OFDM).

By realizing the two-layer signal transmission based on the GTW-OFDM, wherein a low-resolution TV signal is transmitted by the layer A and a high-frequency component is transmitted by the layer B, the vehicle TV receiver can stably receive the low-resolution TV signal regardless of tough ghost. Furthermore, the multi-level signal transmission with respect to the C/N ratio can be realized by differentiating the symbol time t_S based on the CSW-OFDM between the layers A and B. If this CSW-OFDM is combined with the GTW-OFDM, the signal reception in the vehicle TV receiver can be further stabilized. High resolution is not normally required to the vehicle TV or the portable TV.

As the time ratio of the symbol time including a low-resolution TV signal is small, an overall transmission efficiency will not decrease so much even if the guard time is enlarged. Accordingly, using the GTW-OFDM of the present invention for suppressing multipath by laying emphasis on the low-resolution TV signal will realize the multi-layer type TV broadcast service wherein the mobile station such as the portable or vehicle TV receiver can be compatible with the stationary station such as the home TV without substantially lowering the transmission efficiency. If combined with the CSW-OFDM or the C-CDM as described previously, the multi-layer to the C/N ratio can be also realized. Thus, the signal reception in the mobile station will be further stabilized.

An affection of the multipath will be explained in more detail. In case of multipath 810a, 810b, 810c, and 810d having shorter delay time as shown in FIG. 145(a), the signals of both the first and second layers can be received and therefore the HDTV signal can be demodulated. On the contrary, in case of multipath 811a, 811b, 811c, and 811d having longer delay time as shown in FIG. 145(b), the B signal of the second layer cannot be received since its guard time t_gb is not sufficiently long. However, the A signal of the first layer can be received without being bothered by the multipath since its guard time t_ga is sufficiently long. As described above, the B signal includes the high-frequency component of TV signal. The A signal includes the low-frequency component of TV signal. Accordingly, the vehicle TV can reproduce the LDTV signal. Furthermore, as the symbol time Tsa is set larger than symbol time Tsb, the first layer is strong against deterioration of C/N ratio.

Such a discrimination of the guard time and the symbol time is effective to realize two-dimensional multi-layer signal transmission of the OFDM in a simple manner. If the discrimination of guard time is combined with the C-CDM in the circuit shown in FIG. 123, the multi-layer signal transmission effective against both multipath and deterioration of C/N ratio will be realized.

Next, a specific example will be described below.

The smaller the D/U ratio of the receiving signal becomes, the larger the multipath delay time T_M becomes. Because, the reflected wave increases compared with the direct wave. For example, as shown in FIG. 148, if the D/U ratio is smaller than 30 dB, the delay time T_M exceeds 30 us μs because of increase of the reflected wave. Therefore, as can be understood from FIG. 148, it will become possible to receive the signal even in the worst condition if the Tg is set to be larger than 50 us μs.

Accordingly, as shown in detail in FIGS. 149(a) and 149(b), three groups of first 801a, second 801b, and third 801 c 801c layers are assigned in a 2 ms period of 1 sec TV signal. The guard times 797a, 797b, and 797c, i.e. Tga, Tgb, and Tgc, of these three groups are weighted to be, for example, 50 microsounds microseconds, 5 microsounds microseconds, and 1 microsound microsecond, respectively, as shown in FIG. 149(c). Thus, three-layer signal transmission effective to the multipath will be realized as shown in FIG. 150, wherein three layers 801a, 801b, and 801c are provided.

If the GTW-OFDM is applied to ass the picture quality, it is doubtless

At the same time, the multi-layer signal transmission effective to C/N ratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, a two-dimensional multi-layer signal transmission is realized with respect to the multipath and the C/N ratio as shown in FIG. 151. As described previously, it is possible to combine the CSW-OFDM and the C-CDM of the present invention for preventing the overall transmission efficiency from being lowered. In the first, 1-2, and 1-3 layers 801a, 851a, and 851az, the LDTV grade signal can be stably received by, for example, the vehicle TV receiver subjected to the large multipath T_M and low C/N ratio. In the second and 2-3 layers 801b and 851b, the standard-resolution SDTV grade signal can be received by the fixed or stationary station located, for example, in the fringe of the service area which is generally subjected to the lower C/N ratio and ghost. In the third layer 801c which occupies more than half of the service area, the HDTV grade signal can be received since the C/N ratio is high and the ghost is less because of large direct wave. In this manner, a two-dimensional multi-layer broadcast service effective to both the C/N ratio and the multipath can be realized by the combination of the GTW-OFDM and the C-CDM or the combination of the GTW-OFDM and the CSW-C-CDM in accordance with the present invention. thus Thus, the present invention realizes a two-dimensional, matrix type, multi-layer signal transmission system effective to both the C/N ratio and the 194 multipath, which has not ever been realized by the prior art technologies.

A timing chart of a three level (HDTV, SDTV, LDTV) television signal in a two-dimensional multilevel broadcast of three C/N levels and three multipath levels is shown in FIG. 152. As shown in the figure, the LDTV signal is positioned in slot 796a1 of the first level of level layer A, the level with the greatest resistance to multipath interference; the SDTV synchronization signal, address signal, and other important high priority signals are positioned in slot 796a2, which has the next greatest resistance to multipath interference, and slot 796b1, which has strong resistance to C/N deterioration. The SDTV common signal, i.e., low priority signals, and HDTV high priority signals are positioned in levels 2 and 3 of level B. SDTV, EDTV, HDTV, and other high frequency component television signals are positioned in levels 1, 2, and 3 of level C.

As the resistance to C/N deterioration and multipath interference increases, the transmission rate drops, causing the TV signal resolution to drop, and achieving the three-dimensional graceful degradation effect shown in FIG. 153 and unobtainable with conventional methods. As shown in FIG. 153, the three-dimensional multilevel broadcast structure of the invention is achieved with three parameters: C/N ratio, multipath delay time, and the transmission rate.

The present embodiment has been described using the example of a two-dimensional multilevel broadcast structure obtained by combining GTW-OFDM of the invention with C-CDM of the invention as previously described, or combining GTW-OFDM, CSW-C-CDM, but other two-dimensional multilevel broadcast structures can be obtained by combining GTW-OFDM and power-weighted OFDM, or GTW-OFDM with other C/N ratio multilevel transmission methods.

FIG. 154 is obtained by transmitting the power of carriers 794a, 794c, and 794e with less weighting compared with carriers 794b, 794d, and 794f, achieving a two level power-weighted OFDM. Two levels are obtained by power weighting carriers 795a and 795c, which are perpendicular to carrier 794a, to carriers 795b and 795d. While a total of four levels are obtained, the embodiment having only two levels is shown in FIG. 154. As shown in the figure, because the carrier frequencies are distributed, interference with other analog transmissions on the same frequency band is dispersed, and there is minimal adverse effect.

By using a time positioning varying the time width of guard times 797a, 797b, and 797c for each symbol 796a, 796b, and 796c as shown in FIG. 155, three-level multipath multilevel transmission can be achieved. Using the time positioning shown in FIG. 155, the A-, B-, and C-level data is distributed on the time axis. As a result, even if burst noise produced at a specific time occurs, data destruction can be prevented and the TV signal can be stably demodulated by interleaving the data from the different layers. In particular, by interleaving with the A level data distributed, interference from burst noise generated by the ignition systems of other vehicles can be significantly reduced in mobile TV receivers.

Block diagrams of a specific ECC encoder 744j and a specific ECC decoder 749j 759j are shown in FIG. 160a and FIG. 160b, respectively. FIG. 167 is a block diagram of the deinterleaver 936b. The interleave table 954 processed in the deinterleave RAM 936a of the deinterleaver 936b is shown in FIG. 168a, and interleave distance L1 is shown in FIG. 168b.

Burst noise interference can be reduced by interleaving the data in this way. By using a 4-level VSB, 8-level VSB, or 16-level VSB transmission apparatus as described in embodiments 4, 5, and 6, respectively, and shown in the VSB receiver block diagram (FIG. 161) and the VSB transmitter block diagram (FIG. 162), or by using a QAM or PSK transmission apparatus as described in embodiments 1 and 2, respectively, burst noise interference can be reduced, and television reception with very low noise levels can be achieved in ground station broadcasting.

By using 3-level broadcasting by means of the method shown in FIG. 155, LDTV grade television reception by mobile receivers, including mobile TV receivers in motor vehicles and hand-held portable television sets, can be stabilized because level A has the effect of reducing burst noise interference in addition to multipath interference and C/N ratio deterioration.

The multi-level signal transmission method of the present invention is intended to increase the utilization of frequencies but may be suited for not all the transmission systems since causing some type receivers to be declined in the energy utilization. It is a good idea for use with a satellite communications system for selected subscribers to employ most advanced transmitters and receivers designed for best utilization of applicable frequencies and energy. Such a specific purpose signal transmission system will not be bound by the present invention.

The present invention will be advantageous for use with a satellite or terrestrial broadcast service which is essential to run in the same standards for as long as 50 years. During the service period, the broadcast standards must not be altered but improvements will be provided time to time corresponding to up-to-date technological achievements. Particularly, the energy for signal transmission will surely be increased on any satellite. Each TV station should provide a compatible service for guaranteeing TV program signal reception to any type receivers ranging from today's common ones to future advanced ones. The signal transmission system of the present invention can provide a compatible broadcast service of both the existing NTSC and HDTV systems and also, ensure a future extension to match mass date data transmission.

The present invention concerns much on the frequency utilization than the energy utilization. The signal receiving sensitivity of each receiver is arranged different differently depending on a signal state level to be received so that the transmitting power of a transmitter needs not be increased largely. Hence, existing satellites which offer a small energy for reception and transmission of a signal can best be used with the system of the present invention. The system is also arranged for performing the same standards corresponding to an increase in the transmission energy in the future and offering the compatibility between old and new type receivers. In addition, the present invention will be more advantageous for use with the satellite broadcast standards.

The multi-level signal transmission method of the present invention is more preferably employed for terrestrial TV broadcast service in which the energy utilization is not crucial, as compared with satellite broadcast service. The results are such that the signal attenuating regions in a service area which are attributed to a conventional digital HDTV broadcast system are considerably reduced in extension and also, the compatibility of an HDTV receiver or display with the existing NTSC system is obtained. Furthermore, the service area is substantially increased so that program suppliers and sponsors can appreciate more viewers. Although the embodiments of the present invention refer to 16 and 32 QAM procedures, other modulation techniques including 64, 128, and 256 QAM will be employed with equal success. Also, multiple PSK, ASK, and FSK techniques will be applicable as described with the embodiments.

A combination of the TDM with the SRQAM of the present invention has been described in the above. However, the SRQAM of the present invention can be combined also with any of the FDM, CDMA and frequency dispersal communications systems.

Claims

1. A signal transmission and reception apparatus for transmitting and receiving an n-level vsb signal, the apparatus comprising a transmitter and a receiver;

said transmitter comprising:

a compression means for compressing an input video signal to a digital video compression signal;

an error correction encoding means for adding an error correction code to the digital video compression signal to produce an error correction coded signal;

a modulation means for modulating the error correction coded signal to an n-level vsb modulation signal, said modulation means comprising a means for allocating code points along a uniaxial modulation coordinate system, and a filter means having a plurality of coefficients which are a series of impulse responses defined by plotting timebase responses to the vsb modulation signal along the in-phase axis and its orthogonal axis for filtering a series of said code points allocated along the uniaxial modulation coordinate system; and

a transmission means for transmitting the modulation signal, and

said receiver comprising:

a means for receiving a transmitted n-level vsb modulation signal;

a demodulation means for demodulating the received n-level vsb modulation signal into a digital reception signal;

an error correction means for error correcting the digital reception signal to obtain an error-corrected digital signal; and

an expanding means for expanding the error-corrected digital signal to obtain a video output signal.

2. A transmission and reception apparatus according to claim 1, wherein the error correction means comprises a trellis decoder.

3. A transmission and reception apparatus according to claim 2, wherein the trellis decoder is associated with a plurality of memories which each holds a number of selectable correct codes.

4. A transmission and reception apparatus according to claim 1, wherein the digital reception signal is divided into a high priority signal and a low priority signal, and wherein said error correction means comprises a high code gain first error correction means and a low code gain second error correction means, said first error correction means correcting the high priority signal.

5. A transmission and reception apparatus according to claim 4, wherein the high priority signal carries the address data for all data.

6. A transmission and reception apparatus according to claim 4, wherein the first error correction means comprises a trellis decoder.

7. A signal transmission and reception apparatus according to claim 1, further comprising a band path filtering means for filtering the n-level vsb modulation signal before being transmitted.

8. A signal transmission and reception apparatus for transmitting an n-level vsb signal, comprising:

a compression means for compressing an input video signal into a digital video compression signal;

an error correction encoding means for adding an error correction code to the digital video compression signal to produce an error correction coded signal;

a modulation means for modulating the error correction coded signal to an n-level vsb modulation signal, said modulation means comprising a means for allocating code points along a uniaxial modulation coordinate system, and a filter means having a plurality of coefficients which are a series of impulse responses defined by plotting timebase responses to the vsb modulation signal along the in-phase axis and its orthogonal axis for filtering a series of said code points allocated along the uniaxial modulation coordinate system; and

a transmission means for transmitting the modulation signal.

9. A signal transmission apparatus according to claim 8, further comprising a band path filtering means for filtering the n-level vsb modulation signal before being transmitted.

10. A signal receiving apparatus comprising:

a tuner for receiving a transmission signal containing a digital modulation signal and an analog modulation signal and for selecting the digital modulation signal using a local oscillation signal;

an interference detecting means for detecting interference caused by the analog modulation signal from the digital modulation signal selected by the tuner;

a notch filter means responsive to the interference detected by the interference detecting means for removing a carrier of the analog modulation signal in a same frequency band as a frequency band of the digital modulation signal;

an error ratio calculating means for calculating a bit error ratio of an output of the notch filter means; and

an automatic frequency correcting means for changing a frequency of the local oscillation signal of the tuner according to a level of the interference detected by the interference detecting means and the bit error ratio calculated by the error ratio calculating means to compensate for a frequency offset of the carrier of the analog modulated signal.

11. A signal receiving apparatus according to claim 10, wherein the digital modulation signal is an n-level vsb modulation signal.

12. A signal receiving apparatus comprising:

a tuner for receiving a transmission signal containing at least one of a vsb modulated signal and a QAM modulated signal and for selecting one of the vsb modulated signal and the QAM modulated signal to obtain a selected signal;

an analog-to-digital converter for converting the selected signal into a series of digital codes;

a transversal filter provided on an orthogonal axis for suppressing a transmission distortion of the series of digital codes with respect to both orthogonal axes to obtain a series of filtered digital codes allocated on the orthogonal axes;

a carrier recovery means for phase-compensating a carrier of the filtered digital codes allocated on the orthogonal axis outputted from the transversal filter; and

a control means for producing a control signal to extract detected codes at equal time intervals from the vsb modulated signal;

a clock reproducing means for phase synchronizing entire codes of the QAM modulated signal when the selected signal is the QAM modulated signal and for phase synchronizing codes of the vsb modulated signal intermittently at predetermined intervals when the selected signal is the vsb modulated signal; and

a decoding means for decoding an output of the carrier recovery means.

13. A signal transmission apparatus comprising:

a mapper operable to map a first data stream to a 2-level mapped signal having two levels separated by a first amount, and to map a second data stream to an 8-level mapped signal having eight levels, each adjacent two levels of which are separated by a second amount that is less than the first amount;

a dc offset generator operable to add a dc offset to both the 2-level mapped signal and the 8-level mapped signal to produce a 2-level digital signal and an 8-level digital signal, respectively;

a digital filter having an amplitude versus frequency characteristic which comprises (i) a vsb characteristic covering a frequency band including a carrier frequency, and (ii) a roll-off characteristic, said digital filter being operable to filter the 2-level digital signal and the 8-level digital signal to produce a vsb modulated signal; and

a transmitter operable to transmit the vsb modulated signal.

14. A signal receiving apparatus comprising:

a tuner operable to receive a vsb modulated signal resulting from:

(a) a mapper mapping a first data stream to a 2-level mapped signal having two levels separated by a first amount, and a second data stream to an 8-level mapped signal having eight levels, each adjacent two levels of which are separated by the second amount that is less than the first amount;

(b) a dc offset generator adding a dc offset to both the 2-level mapped signal and the 8-level mapped signal to produce a 2-level digital signal and an 8-level digital signal, respectively; and

(c) a digital filter filtering the 2-level digital signal and the 8-level digital signal, wherein the digital filter has an amplitude versus frequency characteristic which comprises (i) a vsb characteristic covering a frequency band including a carrier frequency, and (ii) a roll-off characteristic; and

a demodulator operable to demodulate the received signal to reproduce the first data stream and the second data stream.

15. A signal transmission method comprising:

mapping a first data stream to a 2-level mapped signal having two levels separated by a first amount, and to map a second data stream to an 8-level mapped signal having eight levels, each adjacent two levels of which are separated by a second amount that is less than the first amount;

adding a dc offset to both the 2-level mapped signal and the 8-level mapped signal to produce a 2-level digital signal and an 8-level digital signal, respectively;

filtering the 2-level digital signal and the 8-level digital signal to produce a vsb modulated signal with a digital filter having (i) a vsb characteristic on frequency characteristic covering a frequency band including a carrier frequency, and (ii) a roll-off characteristic; and

transmitting the vsb modulated signal.

16. A signal receiving method comprising:

receiving a vsb modulated signal resulting from:

(a) a mapper mapping a first data stream to an 2-level mapped signal having two levels separated by a first amount, and a second data stream to an 8-level mapped signal having eight levels, each adjacent two levels of which are separated by the second amount that is less than the first amount;

(b) a dc offset generator adding a dc offset to both the 2-level mapped signal and the 8-level mapped signal to produce a 2-level digital signal and an 8-level digital signal, respectively; and

(c) a digital filter filtering the 2-level digital signal and the 8-level digital signal, wherein the digital filter has an amplitude versus frequency characteristic which comprises (i) a vsb characteristic covering a frequency band including a carrier frequency, and (ii) a roll-off characteristic; and

demodulating the received signal to reproduce the first data stream and the second data stream.