A leakage manager system for adequately minimizing static leakage of an integrated circuit is disclosed. The leakage manager system includes a generator configured to generate a control signal to be applied to a sleep transistor. A monitor is configured to determine whether to adjust the control signal to adequately minimize the static leakage. In some embodiments, the monitor includes an emulated sleep transistor. A regulator is configured to adjust the control signal depending on the determination.
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0. 24. An integrated circuit comprising:
a logic component, the logic component being either a logic gate or a storage cell;
a sleep transistor coupled in series to the logic component and to a ground terminal;
a voltage generator configured to generate a negative voltage to be applied to the sleep transistor; and
controller circuitry configured to, while the logic component is in a standby mode:
i) monitor a parameter of an emulated sleep transistor, the parameter of the emulated sleep transistor indicating static leakage of the logic component;
ii) generate a control signal for adjusting the negative voltage to be applied to the sleep transistor based on the parameter of the emulated sleep transistor, thereby reducing static leakage of the integrated circuit.
11. An integrated circuit comprising:
a logic component, the logic component being either a logic gate or a storage cell, and the logic component including;
a sleep transistor in series with the logic component and an electrical connection to a ground terminal;
a voltage generator configured to generate a negative voltage to be applied to the sleep transistor; and
controller circuitry configured to, while the logic component is in a standby mode:
i) induce produce a current through an emulated sleep transistor in proportion to a static leakage current of the integrated circuit;
ii) make a determination of whether to adjust the negative voltage depending on the amount of the current; and
iii) adjust generating a control signal for adjusting the negative voltage depending on the determination.
6. An integrated circuit comprising:
a logic component, the logic component being either a logic gate or a storage cell, and the logic component including;
a sleep transistor in series with the logic component and an electrical connection to a ground terminal;
a voltage generator configured to generate a negative voltage to be applied to the sleep transistor; and
controller circuitry configured to, while the logic component is in a standby mode:
i) monitor a drain-source current and a drain-gate current of either the sleep transistor or through an emulated sleep transistor;
ii) make a determination of whether to adjust the negative voltage in connection with adequate minimization of a static leakage of the integrated circuit; and
iii) adjust generate a control signal for adjusting the negative voltage depending on the determination.
16. An integrated circuit comprising:
a logic component, the logic component being either a logic gate or a storage cell, and the logic component including;
a sleep transistor in series with the logic component and an electrical connection to a ground terminal;
a voltage generator configured to generate a negative voltage to be applied to the sleep transistor; and
a controller, while the logic component is in sleep mode, configured to receive the negative voltage and determine whether to generate a control signal to adjust the negative voltage based on a comparison of a first current and a second current, the controller including:
i) a first emulated sleep transistor configured to receive the negative voltage that defines produces the first current through the first emulated sleep transistor and create a first voltage drop at a drain of the first emulated sleep transistor;
ii) a second emulated sleep transistor configured to receive the negative voltage plus an offset voltage that define produce the second current through the second emulated sleep transistor and create a second voltage drop at a drain of the second emulated sleep transistor; and
iii) circuitry configured to effectively compare the first current to the second current based on a comparison of the first voltage drop and the second voltage drop.
0. 1. An integrated circuit comprising:
a logic component, the logic component being either a logic gate or a storage cell, and the logic component including a sleep transistor in series with an electrical connection to a ground terminal;
a voltage generator configured to generate a negative voltage to be applied to the sleep transistor; and
controller circuitry configured to:
i) monitor drain-source current of the sleep transistor;
ii) make a determination of whether to adjust the negative voltage in connection with adequate minimization of a static leakage of the integrated circuit; and
iii) adjust the negative voltage depending on the determination.
0. 2. The integrated circuit of
0. 3. The integrated circuit of
0. 4. The integrated circuit of
0. 5. The integrated circuit of
7. The integrated circuit of
8. The integrated circuit of
0. 9. The integrated circuit of
12. The integrated circuit of
13. The integrated circuit of
0. 14. The integrated circuit of
0. 17. The integrated circuit of
0. 18. The integrated circuit of
0. 19. The integrated circuit of
0. 20. The integrated circuit of
0. 21. The integrated circuit of
0. 22. The integrated circuit of
0. 25. The integrated circuit of claim 24, wherein the emulated sleep transistor is an NMOS transistor.
0. 26. The integrated circuit of claim 24, wherein the controller circuitry is further configured to continuously make determinations of whether to adjust the negative voltage to reduce the static leakage of the integrated circuit based on the parameter of the emulated sleep transistor.
0. 27. The integrated circuit of claim 24, wherein the controller circuitry is configured to periodically make determinations of whether to adjust the negative voltage to reduce the static leakage of the integrated circuit.
0. 28. The integrated circuit of claim 24, wherein the voltage generator is a charge pump.
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This application is a Continuation of U.S. application Ser. No. 12/620,749, filed Nov. 18, 2009, now U.S. Pat. No. 7,982,532 which is a Divisional of U.S. application Ser. No. 11/998,762, filed Nov. 30, 2007 (now U.S. Pat. No. 7,642,836), which is a Divisional of U.S. application Ser. No. 11/900,971, filed Sep. 14, 2007 (now U.S. Pat. No. 7,382,178), which is a Continuation-in-Part of U.S. application Ser. No. 10/996,739, filed Nov. 24, 2004 (now U.S. Pat. No. 7,279,956), which claims the benefit of U.S. Provisional Application No. 60/586,565, filed Jul. 9, 2004. This application is also related to U.S. application Ser. No. 10/840,893, filed May 7, 2004 (now U.S. Pat. No. 7,051,306).
The entire teachings of the above applications are incorporated herein by reference.
One design goal for integrated circuits is to reduce power consumption. Devices with batteries such as cell phones and laptops particularly need a reduction in power consumption in the integrated circuit to extend the life of the battery. Additionally, a reduction in power consumption prevents over-heating and lowers the heat dissipation of the integrated circuit, which in some cases eliminates or simplifies heat sinks and/or fans required to cool the integrated circuit. As well, the reduction in power consumption of the integrated circuit reduces the AC power draw for the device containing the integrated circuit.
A competing design goal for integrated circuits is increased performance. One way to increase performance is by increasing a maximum operating frequency of a circuit. In order to increase the maximum operating frequency of a circuit, or to integrate more functionality in a smaller area, integrated circuit manufacturing technology shrinks the device size of individual components (e.g. transistors) on the integrated circuit.
However, as component device size scales from 250 nanometers to 130 nanometers or below, a current draw of a device in standby mode referred to as static leakage becomes an increasingly large part of the power budget of the integrated circuit. For example, simulations show that, for an integrated circuit dissipating 50 watts constructed using 130 nanometer devices, greater than 20 percent of the power dissipated is due to static leakage. For even smaller devices, simulations show that the static leakage of an integrated circuit using 50 nanometer feature sizes comprises about 50 percent of the total power budget.
One solution for reducing static leakage includes use of one or more sleep transistors coupled to a logic gate of the integrated circuit. Application of a control signal to the sleep transistor may reduce the static leakage of the logic gate.
A system for minimizing static leakage of an integrated circuit comprises a charge pump, an adaptive leakage controller, and a negative voltage regulator. The charge pump generates a negative voltage to be applied to a sleep transistor. The sleep transistor is configured to control the static leakage of a logic gate of the integrated circuit. In some embodiments, the logic gate may be located in a power island of the integrated circuit. The adaptive leakage controller determines whether to adjust the negative voltage to minimize the static leakage. The adaptive leakage controller may continuously or periodically determine whether to adjust the negative voltage. The negative voltage regulator adjusts the negative voltage depending on the determination.
A method for minimizing static leakage of the integrated circuit comprises generating the negative voltage, applying the negative voltage to the sleep transistor, determining whether to adjust the negative voltage to minimize the static leakage, and adjusting the negative voltage depending on the determination. The method may comprise controlling static leakage of the logic gate of the integrated circuit with the sleep transistor. The method may comprise monitoring one or more parameters of the sleep transistor.
In at least one example embodiment, the adaptive leakage controller determines whether to adjust the negative voltage, and therefore static leakage is minimized with changes in operating temperature of the integrated circuit, or with voltage fluctuations or manufacturing variations. Rather than a fixed negative voltage, the negative voltage applied to the sleep transistor is adjusted to minimize the static leakage. A further advantage is that single threshold transistor circuitry may be utilized in the integrated circuit, reducing the complexity of the manufacturing process for the integrated circuit. A still further advantage is that the negative voltage may be generated within the integrated circuit, obviating components external to the integrated circuit for generating the negative voltage.
According to one example embodiment, there is an integrated circuit that includes two power supply terminals for powering the integrated circuit. The power supply terminals include a Vdd positive supply terminal and a Vss ground terminal together defining a range of logic levels. The integrated circuit also includes logic components. Each of the logic components is a selected one of a logic gate and a storage cell, and each of the logic components includes a sleep transistor in series with each electrical connection to one of the power supply terminals. A voltage generator generates a voltage outside the range of logic levels. The integrated circuit also includes circuitry for applying the voltage outside the range of logic levels to the sleep transistor during a power down mode. In a mode other than power down mode, the circuitry may apply Vdd to the sleep transistor. Alternatively, the circuitry may apply a voltage greater than Vdd to the sleep transistor when in a mode other than power down mode. The integrated circuit also includes a voltage regulator for controlling the voltage generator to adequately minimize leakage current through the sleep transistor during the power down mode.
According to one example embodiment, there is an integrated circuit that includes two power supply terminals for powering the integrated circuit. The power supply terminals include a Vdd positive supply terminal and a Vss ground terminal. The integrated circuit also includes logic components. Each of the logic components is a selected one of a logic gate and a storage cell, and each of the logic components includes a sleep transistor in series with each electrical connection to one of the power supply terminals. A charge pump generates a negative voltage. The integrated circuit also includes circuitry for applying the negative voltage to the sleep transistor during a power down mode. The integrated circuit also includes a voltage regulator for controlling the charge pump to adequately minimize leakage current through the sleep transistor during the power down mode.
According to yet another example embodiment, there is a leakage manager system for adequately minimizing static leakage of an integrated circuit. The leakage manager system includes a generator configured to generate a control signal to be applied to a sleep transistor. A monitor is configured to determine whether to adjust the control signal to adequately minimize the static leakage. The monitor includes an emulated sleep transistor. A regulator is configured to adjust the control signal depending on the determination.
According to yet another example embodiment, there is a method for adequately minimizing static leakage of an integrated circuit having logic components. Each of the logic components is a selected one of a logic gate and a storage cell, and each of the logic components includes a sleep transistor in series with each electrical connection to a Vss ground terminal. The method includes generating a negative voltage to be applied to the sleep transistor. The method also includes determining whether to adjust the negative voltage to adequately minimize the static leakage. The method also includes adjusting the negative voltage depending on the determination.
According to yet another example embodiment, there is an adaptive leakage controller for adequately minimizing a static leakage of an integrated circuit. A capacitor is configured to be charged to a positive supply voltage. A transistor is configured to discharge the capacitor at a rate in proportion to the static leakage. A control circuit is configured to determine whether to adjust a negative voltage applied to a sleep transistor configured to control the static leakage based on a minimum rate of discharge of the capacitor.
According to yet another example embodiment, there is a method for adequately minimizing static leakage of an integrated circuit. The method includes charging a capacitor to a positive supply voltage, and also discharging the capacitor at a rate in proportion to the static leakage. The method also includes adjusting a negative voltage applied to a gate of a sleep transistor to adequately minimize the rate of discharge of the capacitor.
According to yet another example embodiment, there is a power management method carried out in an integrated circuit having logic components, a Vdd positive supply terminal and a Vss ground terminal. Each of the logic components includes a sleep transistor in series with each electrical connection to one of the terminals. The Vdd positive supply terminal and the Vss ground terminal define a range of logic levels. The method includes generating a voltage outside the range of logic levels, and also applying the generated voltage outside the range of logic levels to the sleep transistor during a power down mode. The method also includes adjusting the generated voltage to adequately minimize leakage current through the sleep transistor during the power down mode.
As shown in the exemplary drawings wherein like reference numerals indicate like or corresponding elements among the figures, example embodiments of a system and method according to the present invention are described below in detail. It is to be understood, however, that the present invention may be embodied in various forms. For example, although described herein as pertaining to minimizing static leakage of an integrated circuit, aspects of the invention may be practiced on circuitry not embodied within an integrated circuit. Therefore, specific details disclosed herein are not to be interpreted as limiting, but rather as a basis for the claims and as a representative basis for teaching one skilled in the art to employ the present invention in virtually any appropriately detailed system, structure, method, process or manner.
While
The power island 110 is any section, delineation, partition, or division of the integrated circuit 100 in which power consumption is controlled. In some embodiments, multiple power islands 110 are delineated based on geographical factors of the integrated circuit 100. In some embodiments, multiple power islands 110 are delineated based on functional IP units of the integrated circuit 100. In some embodiments, the power island 110 comprises sub-islands of power to provide further specificity in controlling power in the integrated circuit 100. In some embodiments, each of multiple power islands 110 includes power control circuitry to control power within the power island 110.
The power island manager 120 is any circuitry, device, or system to determine a target power level for one of the power islands 110, determine an action to change a consumption power level of the one of the power islands 110 to the target power level, and perform the action to change the consumption power level of the one of the power islands 110 to the target power level. The power island manager 120 can thus dynamically change the power consumption of the power islands 110 based on the needs and operation of the integrated circuit 100. The target power level is a desired, calculated, or specified power consumption of the power islands 110. The power island manager 120 may be a hierarchy or group of power island managers 120.
While
The power island 110 includes one or more logic gates 115. In an embodiment without the power island 110, the logic gate 115 may comprise any logic gate of the integrated circuit 100. The logic gate 115 of the exemplary embodiment comprises any logic circuitry such as an inverter, a NAND, NOR, exclusive-OR, and exclusive-NOR gate, as well as a storage cells such as a flip-flop and a latch. The logic gate 115 may comprise higher-level Boolean logic, including combinations of individual logic gates.
The logic gate 115 may be powered down to a “sleep mode” in conjunction with a sleep transistor (not shown), as described further herein. To minimize static leakage of the logic gate 115, the leakage manager system 130 generates a negative voltage 150 to be applied to the sleep transistor. Applying the negative voltage 150 to a gate of an NMOS sleep transistor coupled between the logic gate 115 and ground may reduce the static leakage of the logic gate 115. The leakage manager system 130 receives a negative voltage enable signal 140 and subsequently generates and transmits the negative voltage 150 to the power island 110. The negative voltage enable signal 140 may include other signals in addition to the negative voltage enable signal 140. The leakage manager system 130 determines whether to adjust the negative voltage 150. Based on the determination, the leakage manager system 130 adjusts the negative voltage 150, as described further herein.
Adjusting the negative voltage 150 applied to the sleep transistor minimizes static leakage of the logic gate 115. For example, static leakage varies based on parameters such as operating temperature, voltage fluctuations, and manufacturing variations. Therefore, application of a fixed negative voltage to the sleep transistor does not optimally minimize the static leakage of the logic gate 115. Furthermore, generating the negative voltage 150 “on chip” reduces component requirements external to the integrated circuit 100.
An alternative to reduce the static leakage of the logic gate 115 comprises multiple threshold voltage CMOS, in which one or more high threshold transistors are inserted in series with a low threshold logic gate 115. Switching the high threshold transistor “off” reduces the static leakage of the logic gate 115. However, the high threshold transistor requires extra manufacturing process steps for the integrated circuit 100 and slows down the speed of the logic gate 115 as compared to nominal threshold transistors. Providing the negative voltage 150 to a low threshold NMOS sleep transistor advantageously eliminates a requirement to provide high threshold sleep transistor, thereby reducing processing steps needed to manufacture the integrated circuit 100.
As described further herein, the negative voltage regulator 420 of one embodiment generates an enable (EN) signal to the charge pump 430 to enable the charge pump 430 to increase the magnitude of the negative voltage 150 (i.e., to make the negative voltage 150 more negative). If the EN signal is low, an alternating signal from an oscillator 425 to the charge pump 430 is disabled, preventing the charge pump 430 from increasing the magnitude of the negative voltage 150. Alternatively, if the EN signal is high, the alternating signal from the oscillator 425 is enabled so that the charge pump 430 will increase the magnitude of the negative voltage 150. Because the negative voltage regulator 420 toggles the EN signal on or off depending on whether the ALC 410 determines to adjust the negative voltage 150, the leakage manager system 130 maintains the negative voltage 150 at a particular negative voltage to minimize static leakage of the logic gate 115.
At step 530, the ALC 410 determines whether to adjust the negative voltage 150 to minimize static leakage. If the ALC 410 determines to adjust the negative voltage 150, the ALC 410 generates the CTRL signal to the negative voltage regulator 420 (
In one embodiment, the negative voltage regulator 420 continuously adjusts the negative voltage 150. In another embodiment, the negative voltage regulator 420 periodically adjusts the negative voltage 150.
The leakage manager system 130 adjusts the negative voltage 150 to minimize the static leakage of the logic gate 115, even if the static leakage varies due to effects such as temperature variation, voltage fluctuation, or manufacturing process variation. The leakage manager system 130 may advantageously be wholly integrated into the integrated circuit 100, obviating components external to the integrated circuit 100 to generate the negative voltage 150. Further, the leakage manager system 130 may advantageously be utilized in the integrated circuit 100 comprising single threshold transistor logic, so that manufacturing of the integrated circuit 100 is simplified.
It will also be appreciated that although
In
With respect to the second emulated sleep transistor 620, the resistance of the voltage offset transistor 650 reduces the magnitude of the negative voltage 150 (SLPB) by a voltage offset. A gate of the second emulated sleep transistor 620 receives the negative voltage 150 plus the voltage offset. The negative voltage 150 plus the voltage offset produces a second current through the second emulated sleep transistor 620. The second current may comprise drain-gate current and/or drain-source current. The second current creates a second voltage drop across the bias transistors (resistors) 640 at a drain of the second emulated sleep transistor 620. The second voltage drop is sensed at a positive input of the differential amplifier 630.
In operation, the gate of the second emulated sleep transistor 620 operates at a slight voltage offset as compared to the gate of the first emulated sleep transistor 610, because of the voltage offset transistor 650. Referring to
In principle of operation with respect to
In conjunction with the negative voltage regulator 420 of
In this embodiment of the ALC 410, the maximum discharge time for the capacitor 715 corresponding to the lowest value of static leakage is used to generate a digital value for the CTRL signal to the negative voltage regulator 420 (
At step 805, the CTRL signal is initialized to its minimum value. Setting the CTRL signal to its minimum value directs the negative voltage regulator 420 to drive the magnitude of the sleep signal SLPB 150 to its minimum value. At step 810, the controller switches the charging transistor 710 so that the capacitor 715 is charged to VDD. At step 815, the charging transistor 710 is switched off so that the capacitor 715 may discharge through the emulated sleep transistor 720. At step 820, the reference voltage VREF is set to a constant voltage which is less than VDD (e.g. VDD/2). At step 825, the comparator 730 generates an output to the counter 740 after the capacitor 715 discharges to VREF. The counter 740 determines a time required to discharge the capacitor 715 to VREF. The register 750 stores a count (i.e., time) of the counter 740.
At step 827, the CTRL signal is incremented by one bit. At step 830, the controller switches the charging transistor 710 so that the capacitor 715 is again charged to VDD. At step 840, the charging transistor 710 is switched off. At step 860, the comparator 730 generates an output to the counter 740 after the capacitor 715 discharges to VREF. The counter 740 determines the time required to discharge the capacitor 715 with the new value of the CTRL signal and the corresponding SLPB signal.
At step 870, the state logic machine compares the value of the register 750 for the current pass through steps 830-860 (i.e., the time required to discharge the capacitor 715 to VREF for the new value of the CTRL signal and the SLPB signal) to the value of the register 750 for the previous pass through steps 830-860. If the value of the register 750 for the current pass did not decrease relative to the value of the register 750 for the previous pass, then the new value of the CTRL signal corresponds to a lower value of static leakage through the emulated sleep transistor 720. In this case, the method returns to step 827 to further increment the CTRL signal and measure the time required to discharge the capacitor 715. Alternatively, at step 870, if the time required to discharge the capacitor 715 decreased in the current pass, corresponding to a higher value of static leakage through the emulated sleep transistor 720, then the previously stored value of the register 750 corresponds to the lowest value of static leakage through the emulated sleep transistor 720. The value of the CTRL signal corresponding to minimal static leakage is used to control the negative voltage regulator 420 to generate the appropriate setting for the negative voltage 150.
One advantage of the embodiment of the digital ALC 410 of
In an embodiment in conjunction with the analog CTRL signal generated by the ALC 410 of
In conjunction with the digital ALC 410 of
In operation, the negative voltage regulator 420 adjusts the negative voltage 150 depending on a comparison between the fixed voltage reference (point C) and the variable voltage reference (point D). The comparator 920 may generate an enable (EN) signal to enable the charge pump 430 (
The charge pump 430 comprises two interfaces for voltage (e.g., VDD line 1002 and VSS line 1004), an input for an alternating signal (i.e., an INP line 1006), an input for an inverted alternating signal (i.e., an INN line 1008), an inverter 1010, a pump capacitor 1012, capacitances 1014 and 1016, a cross-coupled pass gate 1018 and 1020, PMOS transistors 1022 and 1024, node 1026, an SLP line 1028, an inverter 1030, and an SLPB line 1032. The cross-coupled pass gate 1018 may comprise two PMOS transistors 1038 and 1040. The cross-coupled pass gate 1020 may comprise two PMOS transistors 1042 and 11044 1044. The inverter 1010 may comprise a NMOS transistor 1034 and a PMOS transistor 1036.
In example embodiments, the capacitance 1014 is electrically coupled to INP line 1006 and the capacitance 1016 is electrically coupled to the INN line 1008. The capacitance 1014 and 1016 may comprise a capacitor such as a metal-metal capacitor. In other embodiments, the capacitance 1014 and 1016 may comprise PMOS capacitances (e.g., varactors). Alternately, the capacitance 1014 and 1016 may comprise similar or different components. Those skilled in the art will appreciate that the capacitance 1014 and 1016 may be many different components comprising capacitances. In various embodiments, the capacitances 1014 and 1016 function to smooth out transients from the INP signals and the INN signals, respectively.
The gate of PMOS transistors 1022 and 1024 may be electrically coupled to the capacitance 1014 and 1016, respectively. The PMOS transistor 1022 and PMOS transistor 1024 may be electrically coupled to the pump capacitor 1012. The PMOS transistor 1022 may also be electrically coupled to the NMOS transistor 1034 within inverter 1010 as well as the VSS line 1004, the gate of the PMOS transistor 1038 in the cross coupled pass gate 1018, and the gate of the PMOS transistor 1044 in the cross coupled pass gate 1020. PMOS transistor 1024 may be coupled to SLPB line 1032. In various embodiments, the substrates of PMOS transistor 1022 and 1024 are electrically coupled to node 1026.
The output of the inverter 1010 is electrically coupled to the pump capacitor 1012. The drain of PMOS transistor 1036 is coupled to the source of NMOS transistor 1034 as well as the pump capacitor 1012. The INP line 1006 is electrically coupled to the gates of both the PMOS transistor 1036 and the NMOS transistor 1034 (e.g., the INP line 1006 is electrically coupled to the input of the inverter 1010).
The cross-coupled pass gate 1018 may comprise two PMOS transistors 1038 and 1040. In one example, the PMOS transistor 1038 is electrically coupled to the capacitance 1014, the gate of PMOS transistor 1022, the PMOS transistor 1040, and the gate of PMOS transistor 1042 in the cross-coupled pass gate 1020. The substrate and drain of PMOS transistor 1038 may be electrically coupled to the substrate and drain of the PMOS transistor 1040 as well as the node 1026. The gate of PMOS transistor 1040 is electrically coupled to the PMOS transistors 1042 and 1044 as well as the capacitance 1016 and the gate of PMOS transistor 1024.
The cross-coupled pass gate 1020 may comprise two PMOS transistors 1042 and 1044. In one example, the substrate of the PMOS transistor 1042 is electrically coupled to the substrate of PMOS transistor 1044 and the node 1026. The PMOS transistor 1042 and the PMOS transistor 1044 are electrically coupled to the node 1026.
The cross-coupled pass gate 1018 of this embodiment may be capacitively coupled to the alternating signal (the INP signal) from the oscillator 425 (
The VDD line 1002, VSS line 1004, INP line 1006, INN line 1008, and SLPB line 1032, and SLP line 1028 may comprise wires, traces, or any conductive material configured to function as an electrical medium. The INP line 1006 may be coupled with the oscillator 425 which may generate an alternating signal (i.e., the INP signal). The INN line 1008 may be coupled with an inverter configured to invert the alternating signal (i.e., the INP signal) to generate a complement of the alternating signal. It will be appreciated by those skilled in the art that, in some embodiments, the INN line 1008 receives an alternating signal and the INP line 1006 receives the complement of the alternating signal. There may be many ways to generate the alternating signal and/or the complement of the alternating signal.
Further, the SLPB line 1032 may receive the sleep signal from the leakage manager system 130. In various embodiments, the sleep signal is a negative voltage signal and the SLPB line 1032 is a negative voltage line. The SLP line 1028 may receive the SLP signal (e.g., the enable (EN) signal) from the negative voltage regulator 420. There may be many ways in which the SLP signal may be generated. Further, the SLP signal may be generated in such a way as to make the inversion of the signal either optional or unnecessary (i.e., the inverter 1030 may be optional).
In various embodiments, the alternating signal (INP signal) and the complement of the alternating signal (INN signal) may each comprise two states discussed herein including “high” and “low.” Those skilled in the art will appreciate that the “high” signal is “high” when compared to the “low” state of the signal and is not “high” or “low” in comparison with another standard. In one example, the high state is 1 volt and the low state is 0 or −1 volts. As used herein, the high state is referred to as “high” and the low state is referred to as “low.”
In various embodiments, when the INP signal is low (or goes low), the charge within the pump capacitor 1012 is released through the VSS signal (via VSS line 1004). In one example, the INP signal is received over the INP line 1006 by the gates of the inverter 1010 (i.e., the gate of the PMOS transistor 1036 and the gate of the NMOS transistor 1034). When the INP signal is low (or goes to low), the VDD signal may pass through from the source of the PMOS transistor 1036 to the pump capacitor 1012. Similarly, the INP signal is received by capacitance 1014 and, subsequently, the gate of PMOS transistor 1022. As a result, the charge of the pump capacitor 1012 may be released through the PMOS transistor 1022 and out through the VSS line 1004. The alternate of the INP signal, the INN signal, which is high (or goes to high), is coupled to the capacitance 1016 over the INN line 1008. The gate of PMOS transistor 1024 may receive the high signal from the capacitance 1016. As a result, the PMOS transistor 1024 may decouple the SLPB line 1032 from the pump capacitor 1012.
When the INP signal is high (or goes high), the pump capacitor 1012 is charged (i.e., the capacitor is charged by receiving the VSS signal and the SLPB signal). When the INP signal is high (or goes to high), the PMOS transistor 1036 no longer allows the pump capacitor 1012 to receive the VDD signal. The gate of NMOS transistor 1034 receives the INP signal over the INP line 1006 which subsequently allows the pump capacitor 1012 to receive the VSS signal from VSS line 1004 (the INP signal (i.e., high or going to high) is received by the gate of the PMOS transistor 1022 which prevents the VSS signal from flowing through the PMOS transistor 1022). The alternate of the INP signal, the INN signal (i.e., which is low or goes to low) is received by the gate of PMOS transistor 1024 which subsequently allows the SLPB signal (via the SLPB line 1032) to be received by the pump capacitor 1012 thereby allowing the pump capacitor 1012 to charge.
In some embodiments, the node 1026 is simply tied to ground. In other embodiments, the node 1026 is not tied to ground, but is coupled to the SLP signal. In one example, the SLP signal (via the SLP line 1028) is electrically coupled to the input of inverter 1060, the output of which is coupled to the node 1026. The inverter 1030 may be activated on exiting the sleep mode to prevent a power supply that generates VDD from being shorted to ground through the PMOS transistors 1022 and 1024, and may ensure that any P-N junctions in the wells are not forward biased.
In various embodiments, there is no current flow from the PMOS transistors to the substrate, since the substrate may be at an equal or higher potential than the source and drain of the PMOS transistors. In one example, current flow from the PMOS transistors to the substrate is avoided in order to compete against forward biased diodes for current flow. In another example, to ensure that no P-N junctions in the wells of the PMOS transistors are forward biased, the inverter 1030 may output a complement of the activated SLP signal to drive the node 1026 to 0 V.
The SLP signal may disable the charge pump 430. In one example, the SLP signal (e.g., the EN signal in
Those skilled in art will appreciate that when either the INP signal or the INN signal is high (or goes to high), the signal may electrically couple to the node 1026, in various embodiments. In one example, the INP signal is high and the INN signal is low. The low signal (via the INN line 1008 and the capacitance 1016) is received at the gate of PMOS transistor 1040 which may allow the high INP signal to flow through the PMOS transistor 1040 to the node 1026. In another example, the INN signal is high and the INP signal is low. The low signal (via the INP line 1006 and the capacitance 1014) is received at the gate of PMOS transistor 1042 which may allow the high INN signal to flow through the PMOS transistor 1042 to the node 1026.
In various embodiments, the alternating connectivity of high signals with the node 1026 allows the high signal current to drain at ground when ground is coupled to node 1026. Alternatively, the alternating connectivity of high signals with the node 1026 may electrically couple with the SLP line 1028. In one example, the alternating high signals received by the body of PMOS transistors 1022, 1038, 1040, 1042, 1044, and 1024 (via node 1026) prevent leakage from the pump capacitor 1012 or prevents the VDD signal from coupling to ground. In another example, the alternating high signal over the node 1026 may reduce the voltage required by the SLP signal to sufficiently bias the bodies (i.e., substrates) of the PMOS transistors 1022, 1038, 1040, 1042, 1044, and 1024 in order to disable the charge pump 430.
While one skilled in the art should be able to implement and gain the benefits of the charge pump 430 if provided with only the circuits and diagrams of
With reference to
Due to the INN signal becoming ‘1’, there may be a positive voltage at the capacitance 1016 because this terminal receives the INN signal. With a ‘1’ at the first terminal of the capacitance 1016, there is a ‘0’ at the second terminal of the capacitance 1016. In various embodiments, the gate of the PMOS transistor 1040 and the second terminal of the capacitance 1016 share the same node, so the PMOS transistor 1040 is non-conducting because the gate-to-source voltage difference (VGS) is greater than the threshold voltage (VT). As a result, the cross-coupled pass gate 1018 is non-conducting during the discharging phase. Further, the PMOS transistor 1024 will be off during the discharging phase. As a result, charging of the pump capacitor 1012 does not occur during the discharging phase.
Next is the falling edge of the oscillator 425. In response, the INP signal may go from ‘0’ to ‘1’, and, consequently, the first and second terminals of the capacitance 1014 go from ‘0’ and ‘1’, respectively, to ‘−1’ and ‘0’, respectively. The cross-coupled pass gate 1020 becomes non-conducting because VGS of the transistor 1042 will rise above VT (i.e. the transistor 1042 will become non-conducting).
Also in response to the falling edge of the oscillator 425, the INN signal goes from ‘1’ to ‘0’, and consequently the first and second terminals of the capacitance 1016 go from ‘1’ to ‘0’ and ‘0’ to ‘−1’, respectively. So the node shared by the second terminal of the capacitance 1016, the gate of the second PMOS transistor 1024 and the gate of the PMOS transistor 1040 will be at ‘−1’. The cross-coupled pass gate 1018 will be conducting while a negative voltage is applied at the gate of the PMOS transistor 1040. The second PMOS transistor 1024 is also conducting during this period of time, as the negative voltage is also applied at the gate of the second PMOS transistor 1024. While the second PMOS transistor 1024 is on, the pump capacitor 1012 is charging. The PMOS transistor 1022 may be off during the above-described charging phase, so, in the illustrated example embodiment, discharging of the pump capacitor 1012 does not occur during the charging phase.
With reference now to
Modification of the previously described charge pump 430 to make it suitable for operation in different voltage ranges is contemplated. For example, a higher voltage (for instance, +2V) at the high end of the voltage operation range may be possible by customizing the circuit by switching the INN signal and the INP signal as well as using some bigger circuit components such as, for instance, bigger capacitors.
The components, type of components, and number of components identified in
Further, the above description is illustrative and not restrictive. Many variations of the invention will become apparent to those of skill in the art upon review of this disclosure. The scope of the invention should, therefore, be determined not with reference to the above description, but instead should be determined with reference to the appended claims along with their full scope of equivalents.
Caplan, Randy J., Schwake, Steven J.
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