Apparatus for transmitting broadcast signals, apparatus for receiving broadcast signals, method for transmitting broadcast signals and method for receiving broadcast signals

Abstract

A method and device for processing broadcast signals are discussed. The method includes receiving the broadcast signals carrying service data, demodulating the broadcast signals by an orthogonal frequency division multiplexing (OFDM) scheme, MIMO processing data in the broadcast signals based on a rotation value, where the rotation value has zero (0) degree for a modulation order corresponding to equal to or greater than 64 quadrature amplitude modulation (QAM) and a code rate corresponding to at least one of 2/15, 3/15, 4/15, 5/15, 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15, and the rotation value has zero (0) degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 2/15, 3/15, 4/15 or 5/15, and decoding the data.

Description

CROSS-REFERENCE TO RELATED APPLICATIONS

The LDPC code parameters for PLS1 and PLS2 are as following table 4.

TABLE 4
Sig-
naling				Kldpc			code
Type	Ksig	Kbch	Nbch_parity	(= Nbch)	Nldpc	Nldpc_parity	rate	Qldpc
PLS1	342	1020	60	1080	4320	3240	1/4	36
PLS2	<1021
	>1020	2100		2160	7200	5040	3/10	56

The LDPC parity puncturing block can perform puncturing on the PLS1 data and PLS 2 data.

When shortening is applied to the PLS1 data protection, some LDPC parity bits are punctured after LDPC encoding. Also, for the PLS2 data protection, the LDPC parity bits of PLS2 are punctured after LDPC encoding. These punctured bits are not transmitted.

The bit interleaver 6010 can interleave the each shortened and punctured PLS1 data and PLS2 data.

The constellation mapper 6020 can map the bit interleaved PLS1 data and PLS2 data onto constellations.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 7 illustrates a frame building block according to one embodiment of the present invention.

The frame building block illustrated in FIG. 7 corresponds to an embodiment of the frame building block 1020 described with reference to FIG. 1.

Referring to FIG. 7, the frame building block can include a delay compensation block 7000, a cell mapper 7010 and a frequency interleaver 7020. Description will be given of each block of the frame building block.

The delay compensation block 7000 can adjust the timing between the data pipes and the corresponding PLS data to ensure that they are co-timed at the transmitter end. The PLS data is delayed by the same amount as data pipes are by addressing the delays of data pipes caused by the Input Formatting block and BICM block. The delay of the BICM block is mainly due to the time interleaver 5050. In-band signaling data carries information of the next TI group so that they are carried one frame ahead of the DPs to be signaled. The Delay Compensating block delays in-band signaling data accordingly.

The cell mapper 7010 can map PLS, EAC, FIC, DPs, auxiliary streams and dummy cells into the active carriers of the OFDM symbols in the frame. The basic function of the cell mapper 7010 is to map data cells produced by the TIs for each of the DPs, PLS cells, and EAC/FIC cells, if any, into arrays of active OFDM cells corresponding to each of the OFDM symbols within a frame. Service signaling data (such as PSI (program specific information)/SI) can be separately gathered and sent by a data pipe. The Cell Mapper operates according to the dynamic information produced by the scheduler and the configuration of the frame structure. Details of the frame will be described later.

The frequency interleaver 7020 can randomly interleave data cells received from the cell mapper 7010 to provide frequency diversity. Also, the frequency interleaver 7020 can operate on very OFDM symbol pair comprised of two sequential OFDM symbols using a different interleaving-seed order to get maximum interleaving gain in a single frame.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 8 illustrates an OFDM generation block according to an embodiment of the present invention.

The OFDM generation block illustrated in FIG. 8 corresponds to an embodiment of the OFDM generation block 1030 described with reference to FIG. 1.

The OFDM generation block modulates the OFDM carriers by the cells produced by the Frame Building block, inserts the pilots, and produces the time domain signal for transmission. Also, this block subsequently inserts guard intervals, and applies PAPR (Peak-to-Average Power Radio) reduction processing to produce the final RF signal.

Referring to FIG. 8, the OFDM generation block can include a pilot and reserved tone insertion block 8000, a 2D-eSFN encoding block 8010, an IFFT (Inverse Fast Fourier Transform) block 8020, a PAPR reduction block 8030, a guard interval insertion block 8040, a preamble insertion block 8050, other system insertion block 8060 and a DAC block 8070. Description will be given of each block of the frame building block.

The pilot and reserved tone insertion block 8000 can insert pilots and the reserved tone.

Various cells within the OFDM symbol are modulated with reference information, known as pilots, which have transmitted values known a priori in the receiver. The information of pilot cells is made up of scattered pilots, continual pilots, edge pilots, FSS (frame signaling symbol) pilots and FES (frame edge symbol) pilots. Each pilot is transmitted at a particular boosted power level according to pilot type and pilot pattern. The value of the pilot information is derived from a reference sequence, which is a series of values, one for each transmitted carrier on any given symbol. The pilots can be used for frame synchronization, frequency synchronization, time synchronization, channel estimation, and transmission mode identification, and also can be used to follow the phase noise.

Reference information, taken from the reference sequence, is transmitted in scattered pilot cells in every symbol except the preamble, FSS and FES of the frame. Continual pilots are inserted in every symbol of the frame. The number and location of continual pilots depends on both the FFT size and the scattered pilot pattern. The edge carriers are edge pilots in every symbol except for the preamble symbol. They are inserted in order to allow frequency interpolation up to the edge of the spectrum. FSS pilots are inserted in FSS(s) and FES pilots are inserted in FES. They are inserted in order to allow time interpolation up to the edge of the frame.

The system according to an embodiment of the present invention supports the SFN network, where distributed MISO scheme is optionally used to support very robust transmission mode. The 2D-eSFN is a distributed MISO scheme that uses multiple TX antennas, each of which is located in the different transmitter site in the SFN network.

The 2D-eSFN encoding block 8010 can process a 2D-eSFN processing to distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity in the SFN configuration. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated.

The IFFT block 8020 can modulate the output from the 2D-eSFN encoding block 8010 using OFDM modulation scheme. Any cell in the data symbols which has not been designated as a pilot (or as a reserved tone) carries one of the data cells from the frequency interleaver. The cells are mapped to OFDM carriers.

The PAPR reduction block 8030 can perform a PAPR reduction on input signal using various PAPR reduction algorithms in the time domain.

The guard interval insertion block 8040 can insert guard intervals and the preamble insertion block 8050 can insert preamble in front of the signal. Details of a structure of the preamble will be described later. The other system insertion block 8060 can multiplex signals of a plurality of broadcast transmission/reception systems in the time domain such that data of two or more different broadcast transmission/reception systems providing broadcast services can be simultaneously transmitted in the same RF signal bandwidth. In this case, the two or more different broadcast transmission/reception systems refer to systems providing different broadcast services. The different broadcast services may refer to a terrestrial broadcast service, mobile broadcast service, etc. Data related to respective broadcast services can be transmitted through different frames.

The DAC block 8070 can convert an input digital signal into an analog signal and output the analog signal. The signal output from the DAC block 7800 can be transmitted through multiple output antennas according to the physical layer profiles. A Tx antenna according to an embodiment of the present invention can have vertical or horizontal polarity.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 9 illustrates a structure of an apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention.

The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can correspond to the apparatus for transmitting broadcast signals for future broadcast services, described with reference to FIG. 1.

The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can include a synchronization & demodulation module 9000, a frame parsing module 9010, a demapping & decoding module 9020, an output processor 9030 and a signaling decoding module 9040. A description will be given of operation of each module of the apparatus for receiving broadcast signals.

The synchronization & demodulation module 9000 can receive input signals through m Rx antennas, perform signal detection and synchronization with respect to a system corresponding to the apparatus for receiving broadcast signals and carry out demodulation corresponding to a reverse procedure of the procedure performed by the apparatus for transmitting broadcast signals.

The frame parsing module 9010 can parse input signal frames and extract data through which a service selected by a user is transmitted. If the apparatus for transmitting broadcast signals performs interleaving, the frame parsing module 9010 can carry out deinterleaving corresponding to a reverse procedure of interleaving. In this case, the positions of a signal and data that need to be extracted can be obtained by decoding data output from the signaling decoding module 9040 to restore scheduling information generated by the apparatus for transmitting broadcast signals.

The demapping & decoding module 9020 can convert the input signals into bit domain data and then deinterleave the same as necessary. The demapping & decoding module 9020 can perform demapping for mapping applied for transmission efficiency and correct an error generated on a transmission channel through decoding. In this case, the demapping & decoding module 9020 can obtain transmission parameters necessary for demapping and decoding by decoding the data output from the signaling decoding module 9040.

The output processor 9030 can perform reverse procedures of various compression/signal processing procedures which are applied by the apparatus for transmitting broadcast signals to improve transmission efficiency. In this case, the output processor 9030 can acquire necessary control information from data output from the signaling decoding module 9040. The output of the output processor 8300 corresponds to a signal input to the apparatus for transmitting broadcast signals and may be MPEG-TSs, IP streams (v4 or v6) and generic streams.

The signaling decoding module 9040 can obtain PLS information from the signal demodulated by the synchronization & demodulation module 9000. As described above, the frame parsing module 9010, demapping & decoding module 9020 and output processor 9030 can execute functions thereof using the data output from the signaling decoding module 9040.

FIG. 10, including (a)-(d), illustrates a frame structure according to an embodiment of the present invention.

FIG. 10 shows an example configuration of the frame types and FRUs in a super-frame. FIG. 10(a) shows a super frame according to an embodiment of the present invention, FIG. 10(b) shows FRU (Frame Repetition Unit) according to an embodiment of the present invention, FIG. 10(c) shows frames of variable PHY profiles in the FRU, and FIG. 10(d) shows a structure of a frame.

A super-frame may be composed of eight FRUs. The FRU is a basic multiplexing unit for TDM of the frames, and is repeated eight times in a super-frame.

Each frame in the FRU belongs to one of the PHY profiles, (base, handheld, advanced) or 1-BF. The maximum allowed number of the frames in the FRU is four and a given PHY profile can appear any number of times from zero times to four times in the FRU (e.g., base, base, handheld, advanced). PHY profile definitions can be extended using reserved values of the PHY_PROFILE in the preamble, if required.

The FEF part is inserted at the end of the FRU, if included. When the FEF is included in the FRU, the minimum number of FEFs is 8 in a super-frame. It is not recommended that FBF parts be adjacent to each other.

One frame is further divided into a number of OFDM symbols and a preamble. As shown in (d), the frame comprises a preamble, one or more frame signaling symbols (FSS), normal data symbols and a frame edge symbol (FES).

The preamble is a special symbol that enables fast Futurecast UTB system signal detection and provides a set of basic transmission parameters for efficient transmission and reception of the signal. The detailed description of the preamble will be will be described later.

The main purpose of the FSS(s) is to carry the PLS data. For fast synchronization and channel estimation, and hence fast decoding of PLS data, the FSS has more dense pilot pattern than the normal data symbol. The FES has exactly the same pilots as the FSS, which enables frequency-only interpolation within the FES and temporal interpolation, without extrapolation, for symbols immediately preceding the FES.

FIG. 11 illustrates a signaling hierarchy structure of the frame according to an embodiment of the present invention.

FIG. 11 illustrates the signaling hierarchy structure, which is split into three main parts: the preamble signaling data 11000, the PLS1 data 11010 and the PLS2 data 11020. The purpose of the preamble, which is carried by the preamble symbol in every frame, is to indicate the transmission type and basic transmission parameters of that frame. The PLS1 enables the receiver to access and decode the PLS2 data, which contains the parameters to access the DP of interest. The PLS2 is carried in every frame and split into two main parts: PLS2-STAT data and PLS2-DYN data. The static and dynamic portion of PLS2 data is followed by padding, if necessary.

FIG. 12 illustrates preamble signaling data according to an embodiment of the present invention.

Preamble signaling data carries 21 bits of information that are needed to enable the receiver to access PLS data and trace DPs within the frame structure. Details of the preamble signaling data are as follows:

PHY_PROFILE: This 3-bit field indicates the PHY profile type of the current frame. The mapping of different PHY profile types is given in below table 5.

TABLE 5

Value   PHY profile

000     Base profile
001     Handheld profile
010     Advanced profiled
011~110 Reserved
111     FEF

FFT_SIZE: This 2 bit field indicates the FFT size of the current frame within a frame-group, as described in below table 6.

TABLE 6

Value FFT size

00    8K FFT
01    16K FFT
10    32K FFT
11    Reserved

GI_FRACTION: This 3 bit field indicates the guard interval fraction value in the current super-frame, as described in below table 7.

TABLE 7

Value   GI_FRACTION

000     ⅕
001     1/10
010     1/20
011     1/40
100     1/80
101     1/160
110~111 Reserved

EAC_FLAG: This 1 bit field indicates whether the EAC is provided in the current frame. If this field is set to ‘1’, emergency alert service (EAS) is provided in the current frame. If this field set to ‘0’, EAS is not carried in the current frame. This field can be switched dynamically within a super-frame.

PILOT_MODE: This 1-bit field indicates whether the pilot mode is mobile mode or fixed mode for the current frame in the current frame-group. If this field is set to ‘0’, mobile pilot mode is used. If the field is set to ‘1’, the fixed pilot mode is used.

PAPR_FLAG: This 1-bit field indicates whether PAPR reduction is used for the current frame in the current frame-group. If this field is set to value ‘1’, tone reservation is used for PAPR reduction. If this field is set to ‘0’, PAPR reduction is not used.

FRU_CONFIGURE: This 3-bit field indicates the PHY profile type configurations of the frame repetition units (FRU) that are present in the current super-frame. All profile types conveyed in the current super-frame are identified in this field in all preambles in the current super-frame. The 3-bit field has a different definition for each profile, as show in below table 8.

TABLE 8
	Current	Current	Current
	PHY_	PHY_	PHY_	Current
	PROFILE =	PROFILE = ‘001’	PROFILE = ‘010’	PHY_PROFILE =
	‘000’ (base)	(handheld)	(advanced)	‘111’ (FEF)
FRU_CONFIGURE =	Only base	Only handheld	Only advanced	Only FEF
000	profile	profile present	profile present	present
	present
FRU_CONFIGURE =	Handheld	Base profile	Base profile	Base profile
1XX	profile present	present	present	present
FRU_CONFIGURE =	Advanced	Advanced	Handheld	Handheld
X1X	profile	profile	profile	profile
	present	present	present	present
FRU_CONFIGURE =	FEF	FEF	FEF	Advanced
XX1	present	present	present	profile
				present

RESERVED: This 7-bit field is reserved for future use.

FIG. 13 illustrates PLS1 data according to an embodiment of the present invention.

PLS1 data provides basic transmission parameters including parameters required to enable the reception and decoding of the PLS2. As above mentioned, the PLS1 data remain unchanged for the entire duration of one frame-group. The detailed definition of the signaling fields of the PLS1 data are as follows:

PREAMBLE_DATA: This 20-bit field is a copy of the preamble signaling data excluding the EAC_FLAG.

NUM_FRAME_FRU: This 2-bit field indicates the number of the frames per FRU.

PAYLOAD_TYPE: This 3-bit field indicates the format of the payload data carried in the frame-group. PAYLOAD_TYPE is signaled as shown in table 9.

TABLE 9

value Payload type

1XX   TS stream is transmitted
X1X   IP stream is transmitted
XX1   GS stream is transmitted

NUM_FSS: This 2-bit field indicates the number of FSS symbols in the current frame.

SYSTEM_VERSION: This 8-bit field indicates the version of the transmitted signal format. The SYSTEM_VERSION is divided into two 4-bit fields, which are a major version and a minor version.

Major version: The MSB four bits of SYSTEM_VERSION field indicate major version information. A change in the major version field indicates a non-backward-compatible change. The default value is ‘0000’. For the version described in this standard, the value is set to ‘0000’.

Minor version: The LSB four bits of SYSTEM_VERSION field indicate minor version information. A change in the minor version field is backward-compatible.

CELL_ID: This is a 16-bit field which uniquely identifies a geographic cell in an ATSC network. An ATSC cell coverage area may consist of one or more frequencies, depending on the number of frequencies used per Futurecast UTB system. If the value of the CELL_ID is not known or unspecified, this field is set to ‘0’.

NETWORK_ID: This is a 16-bit field which uniquely identifies the current ATSC network.

SYSTEM_ID: This 16-bit field uniquely identifies the Futurecast UTB system within the ATSC network. The Futurecast UTB system is the terrestrial broadcast system whose input is one or more input streams (TS, IP, GS) and whose output is an RF signal. The Futurecast UTB system carries one or more PHY profiles and FEF, if any. The same Futurecast UTB system may carry different input streams and use different RF frequencies in different geographical areas, allowing local service insertion. The frame structure and scheduling are controlled in one place and is identical for all transmissions within a Futurecast UTB system. One or more Futurecast UTB systems may have the same SYSTEM_ID meaning that they all have the same physical layer structure and configuration.

The following loop consists of FRU_PHY_PROFILE, FRU_FRAME_LENGTH, FRU_GI_FRACTION, and RESERVED which are used to indicate the FRU configuration and the length of each frame type. The loop size is fixed so that four PHY profiles (including a FEF) are signaled within the FRU. If NUM_FRAME_FRU is less than 4, the unused fields are filled with zeros.

FRU_PHY_PROFILE: This 3-bit field indicates the PHY profile type of the (i+1)th (i is the loop index) frame of the associated FRU. This field uses the same signaling format as shown in the table 8.

FRU_FRAME_LENGTH: This 2-bit field indicates the length of the (i+1)th frame of the associated FRU. Using FRU_FRAME_LENGTH together with FRU_GI_FRACTION, the exact value of the frame duration can be obtained.

FRU_GI_FRACTION: This 3-bit field indicates the guard interval fraction value of the (i+1)th frame of the associated FRU. FRU_GI_FRACTION is signaled according to the table 7.

RESERVED: This 4-bit field is reserved for future use.

The following fields provide parameters for decoding the PLS2 data.

PLS2_FEC_TYPE: This 2-bit field indicates the FEC type used by the PLS2 protection. The FEC type is signaled according to table 10. The details of the LDPC codes will be described later.

TABLE 10

Content PLS2 FEC type

00      4K-1/4 and 7K-3/10 LDPC codes
01~11   Reserved

PLS2_MOD: This 3-bit field indicates the modulation type used by the PLS2. The modulation type is signaled according to table 11.

TABLE 11

Value   PLS2_MODE

000     BPSK
001     QPSK
010     QAM-16
011     NUQ-64
100~111 Reserved

PLS2_SIZE_CELL: This 15-bit field indicates Ctotal_partial_block, the size (specified as the number of QAM cells) of the collection of full coded blocks for PLS2 that is carried in the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_STAT_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-STAT for the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_DYN_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-DYN for the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_REP_FLAG: This 1-bit flag indicates whether the PLS2 repetition mode is used in the current frame-group. When this field is set to value ‘1’, the PLS2 repetition mode is activated. When this field is set to value ‘0’, the PLS2 repetition mode is deactivated.

PLS2_REP_SIZE_CELL: This 15-bit field indicates Ctotal_partial_block, the size (specified as the number of QAM cells) of the collection of partial coded blocks for PLS2 carried in every frame of the current frame-group, when PLS2 repetition is used. If repetition is not used, the value of this field is equal to 0. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_FEC_TYPE: This 2-bit field indicates the FEC type used for PLS2 that is carried in every frame of the next frame-group. The FEC type is signaled according to the table 10.

PLS2_NEXT_MOD: This 3-bit field indicates the modulation type used for PLS2 that is carried in every frame of the next frame-group. The modulation type is signaled according to the table 11.

PLS2_NEXT_REP_FLAG: This 1-bit flag indicates whether the PLS2 repetition mode is used in the next frame-group. When this field is set to value ‘1’, the PLS2 repetition mode is activated. When this field is set to value ‘0’, the PLS2 repetition mode is deactivated.

PLS2_NEXT_REP_SIZE_CELL: This 15-bit field indicates Ctotal_full_block, the size (specified as the number of QAM cells) of the collection of full coded blocks for PLS2 that is carried in every frame of the next frame-group, when PLS2 repetition is used. If repetition is not used in the next frame-group, the value of this field is equal to 0. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_REP_STAT_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-STAT for the next frame-group. This value is constant in the current frame-group.

PLS2_NEXT_REP_DYN_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-DYN for the next frame-group. This value is constant in the current frame-group.

PLS2_AP_MODE: This 2-bit field indicates whether additional parity is provided for PLS2 in the current frame-group. This value is constant during the entire duration of the current frame-group. The below table 12 gives the values of this field. When this field is set to ‘00’, additional parity is not used for the PLS2 in the current frame-group.

TABLE 12

Value PLS2-AP mode

00    AP is not provided
01    AP1 mode
10~11 Reserved

PLS2_AP_SIZE_CELL: This 15-bit field indicates the size (specified as the number of QAM cells) of the additional parity bits of the PLS2. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_AP_MODE: This 2-bit field indicates whether additional parity is provided for PLS2 signaling in every frame of next frame-group. This value is constant during the entire duration of the current frame-group. The table 12 defines the values of this field

PLS2_NEXT_AP_SIZE_CELL: This 15-bit field indicates the size (specified as the number of QAM cells) of the additional parity bits of the PLS2 in every frame of the next frame-group. This value is constant during the entire duration of the current frame-group.

RESERVED: This 32-bit field is reserved for future use.

CRC_32: A 32-bit error detection code, which is applied to the entire PLS1 signaling.

FIG. 14 illustrates PLS2 data according to an embodiment of the present invention.

FIG. 14 illustrates PLS2-STAT data of the PLS2 data. The PLS2-STAT data are the same within a frame-group, while the PLS2-DYN data provide information that is specific for the current frame.

The details of fields of the PLS2-STAT data are as follows:

FIC_FLAG: This 1-bit field indicates whether the FIC is used in the current frame-group. If this field is set to ‘1’, the FIC is provided in the current frame. If this field set to ‘0’, the FIC is not carried in the current frame. This value is constant during the entire duration of the current frame-group.

AUX_FLAG: This 1-bit field indicates whether the auxiliary stream(s) is used in the current frame-group. If this field is set to ‘1’, the auxiliary stream is provided in the current frame. If this field set to ‘0’, the auxiliary stream is not carried in the current frame. This value is constant during the entire duration of current frame-group.

NUM_DP: This 6-bit field indicates the number of DPs carried within the current frame. The value of this field ranges from 1 to 64, and the number of DPs is NUM_DP+1.

DP_ID: This 6-bit field identifies uniquely a DP within a PHY profile.

DP_TYPE: This 3-bit field indicates the type of the DP. This is signaled according to the below table 13.

TABLE 13

Value   DP Type

000     DP Type 1
001     DP Type 2
010~111 reserved

DP_GROUP_ID: This 8-bit field identifies the DP group with which the current DP is associated. This can be used by a receiver to access the DPs of the service components associated with a particular service, which will have the same DP_GROUP_ID.

BASE_DP_ID: This 6-bit field indicates the DP carrying service signaling data (such as PSI/SI) used in the Management layer. The DP indicated by BASE_DP_ID may be either a normal DP carrying the service signaling data along with the service data or a dedicated DP carrying only the service signaling data

DP_FEC_TYPE: This 2-bit field indicates the FEC type used by the associated DP. The FEC type is signaled according to the below table 14.

TABLE 14

Value FEC_TYPE

00    16K LDPC
01    64K LDPC
10~11 Reserved

DP_COD: This 4-bit field indicates the code rate used by the associated DP. The code rate is signaled according to the below table 15.

TABLE 15

Value     Code rate

0000      5/15
0001      6/15
0010      7/15
0011      8/15
0100      9/15
0101      10/15
0110      11/15
0111      12/15
1000      13/15
1001~1111 Reserved

DP_MOD: This 4-bit field indicates the modulation used by the associated DP. The modulation is signaled according to the below table 16.

TABLE 16

Value     Modulation

0000      QPSK
0001      QAM-16
0010      NUQ-64
0011      NUQ-256
0100      NUQ-1024
0101      NUC-16
0110      NUC-64
0111      NUC-256
1000      NUC-1024
1001~1111 reserved

DP_SSD_FLAG: This 1-bit field indicates whether the SSD mode is used in the associated DP. If this field is set to value ‘1’, SSD is used. If this field is set to value ‘0’, SSD is not used.

The following field appears only if PHY_PROFILE is equal to ‘010’, which indicates the advanced profile:

DP_MIMO: This 3-bit field indicates which type of MIMO encoding process is applied to the associated DP. The type of MIMO encoding process is signaled according to the table 17.

TABLE 17

Value   MIMO encoding

000     FR-SM
001     FRFD-SM
010~111 reserved

DP_TI_TYPE: This 1-bit field indicates the type of time-interleaving. A value of ‘0’ indicates that one TI group corresponds to one frame and contains one or more TI-blocks. A value of ‘1’ indicates that one TI group is carried in more than one frame and contains only one TI-block.

DP_TI_LENGTH: The use of this 2-bit field (the allowed values are only 1, 2, 4, 8) is determined by the values set within the DP_TI_TYPE field as follows:

If the DP_TI_TYPE is set to the value ‘1’, this field indicates PI, the number of the frames to which each TI group is mapped, and there is one TI-block per TI group (NTI=1). The allowed PI values with 2-bit field are defined in the below table 18.

If the DP_TI_TYPE is set to the value ‘0’, this field indicates the number of TI-blocks NTI per TI group, and there is one TI group per frame (PI=1). The allowed PI values with 2-bit field are defined in the below table 18.

TABLE 18
2-bit field	PI	NTI
00	1	1
01	2	2
10	4	3
11	8	4

DP_FRAME_INTERVAL: This 2-bit field indicates the frame interval (HUMP) within the frame-group for the associated DP and the allowed values are 1, 2, 4, 8 (the corresponding 2-bit field is ‘00’, ‘01’, ‘10’, or ‘11’, respectively). For DPs that do not appear every frame of the frame-group, the value of this field is equal to the interval between successive frames. For example, if a DP appears on the frames 1, 5, 9, 13, etc., this field is set to ‘4’. For DPs that appear in every frame, this field is set to ‘1’.

DP_TI_BYPASS: This 1-bit field determines the availability of time interleaver 5050. If time interleaving is not used for a DP, it is set to ‘1’. Whereas if time interleaving is used it is set to ‘0’.

DP_FIRST_FRAME_IDX: This 5-bit field indicates the index of the first frame of the super-frame in which the current DP occurs. The value of DP_FIRST_FRAME_IDX ranges from 0 to 31

DP_NUM_BLOCK_MAX: This 10-bit field indicates the maximum value of DP_NUM_BLOCKS for this DP. The value of this field has the same range as DP_NUM_BLOCKS.

DP_PAYLOAD_TYPE: This 2-bit field indicates the type of the payload data carried by the given DP. DP_PAYLOAD_TYPE is signaled according to the below table 19.

TABLE 19

Value Payload Type

00    TS.
01    IP
10    GS
11    reserved

DP_INBAND_MODE: This 2-bit field indicates whether the current DP carries in-band signaling information. The in-band signaling type is signaled according to the below table 20.

TABLE 20

Value In-band mode

00    In-band signaling is not carried.
01    INBAND-PLS is carried only
10    INBAND-ISSY is carried only
11    INBAND-PLS and INBAND-ISSY are carried

DP_PROTOCOL_TYPE: This 2-bit field indicates the protocol type of the payload carried by the given DP. It is signaled according to the below table 21 when input payload types are selected.

TABLE 21
	If DP_	If DP_	If DP_
	PAYLOAD_TYPE	PAYLOAD_TYPE	PAYLOAD_TYPE
Value	Is TS	Is IP	Is GS
00	MPEG2-TS	IPv4	(Note)
01	Reserved	IPv6	Reserved
10	Reserved	Reserved	Reserved
11	Reserved	Reserved	Reserved

DP_CRC_MODE: This 2-bit field indicates whether CRC encoding is used in the Input Formatting block. The CRC mode is signaled according to the below table 22.

TABLE 22

Value CRC mode

00    Not used
01    CRC-8
10    CRC-16
11    CRC-32

DNP_MODE: This 2-bit field indicates the null-packet deletion mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). DNP_MODE is signaled according to the below table 23. If DP_PAYLOAD_TYPE is not TS (‘00’), DNP_MODE is set to the value ‘00’.

TABLE 23

Value Null-packet deletion mode

00    Not used
01    DNP-NORMAL
10    DNP-OFFSET
11    reserved

ISSY_MODE: This 2-bit field indicates the ISSY mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). The ISSY_MODE is signaled according to the below table 24 If DP_PAYLOAD_TYPE is not TS (‘00’), ISSY_MODE is set to the value ‘00’.

TABLE 24

Value ISSY mode

00    Not used
01    ISSY-UP
10    ISSY-BBF
11    reserved

HC_MODE_TS: This 2-bit field indicates the TS header compression mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). The HC_MODE_TS is signaled according to the below table 25.

TABLE 25

Value Header compression mode

00    HC_MODE_TS 1
01    HC_MODE_TS 2
10    HC_MODE_TS 3
11    HC_MODE_TS 4

HC_MODE_IP: This 2-bit field indicates the IP header compression mode when DP_PAYLOAD_TYPE is set to IP (‘01’). The HC_MODE_IP is signaled according to the below table 26.

TABLE 26

Value Header compression mode

00    No compression
01    HC_MODE_IP 1
10~41 reserved

PID: This 13-bit field indicates the PID number for TS header compression when DP_PAYLOAD_TYPE is set to TS (‘00’) and HC_MODE_TS is set to ‘01’ or ‘10’.

RESERVED: This 8-bit field is reserved for future use.

The following field appears only if FIC_FLAG is equal to ‘1’:

FIC_VERSION: This 8-bit field indicates the version number of the FIC.

FIC_LENGTH_BYTE: This 13-bit field indicates the length, in bytes, of the FIC.

RESERVED: This 8-bit field is reserved for future use.

The following field appears only if AUX_FLAG is equal to ‘1’:

NUM_AUX: This 4-bit field indicates the number of auxiliary streams. Zero means no auxiliary streams are used.

AUX_CONFIG_RFU: This 8-bit field is reserved for future use.

AUX_STREAM_TYPE: This 4-bit is reserved for future use for indicating the type of the current auxiliary stream.

AUX_PRIVATE_CONFIG: This 28-bit field is reserved for future use for signaling auxiliary streams.

FIG. 15 illustrates PLS2 data according to another embodiment of the present invention.

FIG. 15 illustrates PLS2-DYN data of the PLS2 data. The values of the PLS2-DYN data may change during the duration of one frame-group, while the size of fields remains constant.

The details of fields of the PLS2-DYN data are as follows:

FRAME_INDEX: This 5-bit field indicates the frame index of the current frame within the super-frame. The index of the first frame of the super-frame is set to ‘0’.

PLS_CHANGE_COUNTER: This 4-bit field indicates the number of super-frames ahead where the configuration will change. The next super-frame with changes in the configuration is indicated by the value signaled within this field. If this field is set to the value ‘0000’, it means that no scheduled change is foreseen: e.g., value ‘1’ indicates that there is a change in the next super-frame.

FIC_CHANGE_COUNTER: This 4-bit field indicates the number of super-frames ahead where the configuration (i.e., the contents of the FIC) will change. The next super-frame with changes in the configuration is indicated by the value signaled within this field. If this field is set to the value ‘0000’, it means that no scheduled change is foreseen: e.g. value ‘0001’ indicates that there is a change in the next super-frame.

RESERVED: This 16-bit field is reserved for future use.

The following fields appear in the loop over NUM_DP, which describe the parameters associated with the DP carried in the current frame.

DP_ID: This 6-bit field indicates uniquely the DP within a PHY profile.

DP_START: This 15-bit (or 13-bit) field indicates the start position of the first of the DPs using the DPU addressing scheme. The DP_START field has differing length according to the PHY profile and FFT size as shown in the below table 27.

TABLE 27
		DP_START field size
	PHY profile	64K	16K
	Base	13 bit	15 bit
	Handheld	—	13 bit
	Advanced	13 bit	15 bit

DP_NUM_BLOCK: This 10-bit field indicates the number of FEC blocks in the current TI group for the current DP. The value of DP_NUM_BLOCK ranges from 0 to 1023

RESERVED: This 8-bit field is reserved for future use.

The following fields indicate the FIC parameters associated with the EAC.

EAC_FLAG: This 1-bit field indicates the existence of the EAC in the current frame. This bit is the same value as the EAC_FLAG in the preamble.

EAS_WAKE_UP_VERSION_NUM: This 8-bit field indicates the version number of a wake-up indication.

If the EAC_FLAG field is equal to ‘1’, the following 12 bits are allocated for EAC_LENGTH_BYTE field. If the EAC_FLAG field is equal to ‘0’, the following 12 bits are allocated for EAC_COUNTER.

EAC_LENGTH_BYTE: This 12-bit field indicates the length, in byte, of the EAC.

EAC_COUNTER: This 12-bit field indicates the number of the frames before the frame where the EAC arrives.

The following field appears only if the AUX_FLAG field is equal to ‘1’:

AUX_PRIVATE_DYN: This 48-bit field is reserved for future use for signaling auxiliary streams. The meaning of this field depends on the value of AUX_STREAM_TYPE in the configurable PLS2-STAT.

CRC_32: A 32-bit error detection code, which is applied to the entire PLS2.

FIG. 16 illustrates a logical structure of a frame according to an embodiment of the present invention.

As above mentioned, the PLS, EAC, FIC, DPs, auxiliary streams and dummy cells are mapped into the active carriers of the OFDM symbols in the frame. The PLS1 and PLS2 are first mapped into one or more FSS(s). After that, EAC cells, if any, are mapped immediately following the PLS field, followed next by FIC cells, if any. The DPs are mapped next after the PLS or EAC, FIC, if any. Type 1 DPs follows first, and Type 2 DPs next. The details of a type of the DP will be described later. In some case, DPs may carry some special data for EAS or service signaling data. The auxiliary stream or streams, if any, follow the DPs, which in turn are followed by dummy cells. Mapping them all together in the above mentioned order, i.e. PLS, EAC, FIC, DPs, auxiliary streams and dummy data cells exactly fill the cell capacity in the frame.

FIG. 17 illustrates PLS mapping according to an embodiment of the present invention.

PLS cells are mapped to the active carriers of FSS(s). Depending on the number of cells occupied by PLS, one or more symbols are designated as FSS(s), and the number of FSS(s) NFSS is signaled by NUM_FSS in PLS1. The FSS is a special symbol for carrying PLS cells. Since robustness and latency are critical issues in the PLS, the FSS(s) has higher density of pilots allowing fast synchronization and frequency-only interpolation within the FSS.

PLS cells are mapped to active carriers of the NFSS FSS(s) in a top-down manner as shown in an example in FIG. 17. The PLS1 cells are mapped first from the first cell of the first FSS in an increasing order of the cell index. The PLS2 cells follow immediately after the last cell of the PLS1 and mapping continues downward until the last cell index of the first FSS. If the total number of required PLS cells exceeds the number of active carriers of one FSS, mapping proceeds to the next FSS and continues in exactly the same manner as the first FSS.

After PLS mapping is completed, DPs are carried next. If EAC, FIC or both are present in the current frame, they are placed between PLS and “normal” DPs.

FIG. 18 illustrates EAC mapping according to an embodiment of the present invention.

EAC is a dedicated channel for carrying EAS messages and links to the DPs for EAS. EAS support is provided but EAC itself may or may not be present in every frame. EAC, if any, is mapped immediately after the PLS2 cells. EAC is not preceded by any of the FIC, DPs, auxiliary streams or dummy cells other than the PLS cells. The procedure of mapping the EAC cells is exactly the same as that of the PLS.

The EAC cells are mapped from the next cell of the PLS2 in increasing order of the cell index as shown in the example in FIG. 18. Depending on the EAS message size, EAC cells may occupy a few symbols, as shown in FIG. 18.

EAC cells follow immediately after the last cell of the PLS2, and mapping continues downward until the last cell index of the last FSS. If the total number of required EAC cells exceeds the number of remaining active carriers of the last FSS mapping proceeds to the next symbol and continues in exactly the same manner as FSS(s). The next symbol for mapping in this case is the normal data symbol, which has more active carriers than a FSS.

After EAC mapping is completed, the FIC is carried next, if any exists. If FIC is not transmitted (as signaled in the PLS2 field), DPs follow immediately after the last cell of the EAC.

FIG. 19, including (a) and (b), illustrates FIC mapping according to an embodiment of the present invention.

FIG. 19(a) shows an example mapping of FIC cell without EAC, and FIG. 19(b) shows an example mapping of FIC cell with EAC.

FIC is a dedicated channel for carrying cross-layer information to enable fast service acquisition and channel scanning. This information primarily includes channel binding information between DPs and the services of each broadcaster. For fast scan, a receiver can decode FIC and obtain information such as broadcaster ID, number of services, and BASE_DP_ID. For fast service acquisition, in addition to FIC, base DP can be decoded using BASE_DP_ID. Other than the content it carries, a base DP is encoded and mapped to a frame in exactly the same way as a normal DP. Therefore, no additional description is required for a base DP. The FIC data is generated and consumed in the Management Layer. The content of FIC data is as described in the Management Layer specification.

The FIC data is optional and the use of FIC is signaled by the FIC_FLAG parameter in the static part of the PLS2. If FIC is used, FIC_FLAG is set to ‘1’ and the signaling field for FIC is defined in the static part of PLS2. Signaled in this field are FIC_VERSION, and FIC_LENGTH_BYTE. FIC uses the same modulation, coding and time interleaving parameters as PLS2. FIC shares the same signaling parameters such as PLS2_MOD and PLS2_FEC. FIC data, if any, is mapped immediately after PLS2 or EAC if any. FIC is not preceded by any normal DPs, auxiliary streams or dummy cells. The method of mapping FIC cells is exactly the same as that of EAC which is again the same as PLS.

Without EAC after PLS, FIC cells are mapped from the next cell of the PLS2 in an increasing order of the cell index as shown in an example in (a). Depending on the FIC data size, FIC cells may be mapped over a few symbols, as shown in (b).

FIC cells follow immediately after the last cell of the PLS2, and mapping continues downward until the last cell index of the last FSS. If the total number of required FIC cells exceeds the number of remaining active carriers of the last FSS, mapping proceeds to the next symbol and continues in exactly the same manner as FSS(s). The next symbol for mapping in this case is the normal data symbol which has more active carriers than a FSS.

If EAS messages are transmitted in the current frame, EAC precedes FIC, and FIC cells are mapped from the next cell of the EAC in an increasing order of the cell index as shown in (b).

After FIC mapping is completed, one or more DPs are mapped, followed by auxiliary streams, if any, and dummy cells.

FIG. 20, including (a) and (b), illustrates a type of DP according to an embodiment of the present invention.

FIG. 20(a) shows type 1 DP, and FIG. 20(b) shows type 2 DP.

After the preceding channels, i.e., PLS, EAC and FIC, are mapped, cells of the DPs are mapped. A DP is categorized into one of two types according to mapping method:

Type 1 DP: DP is mapped by TDM

Type 2 DP: DP is mapped by FDM

The type of DP is indicated by DP_TYPE field in the static part of PLS2. FIG. 20 illustrates the mapping orders of Type 1 DPs and Type 2 DPs. Type 1 DPs are first mapped in the increasing order of cell index, and then after reaching the last cell index, the symbol index is increased by one. Within the next symbol, the DP continues to be mapped in the increasing order of cell index starting from p=0. With a number of DPs mapped together in one frame, each of the Type 1 DPs are grouped in time, similar to TDM multiplexing of DPs.

Type 2 DPs are first mapped in the increasing order of symbol index, and then after reaching the last OFDM symbol of the frame, the cell index increases by one and the symbol index rolls back to the first available symbol and then increases from that symbol index. After mapping a number of DPs together in one frame, each of the Type 2 DPs are grouped in frequency together, similar to FDM multiplexing of DPs.

Type 1 DPs and Type 2 DPs can coexist in a frame if needed with one restriction; Type 1 DPs always precede Type 2 DPs. The total number of OFDM cells carrying Type 1 and Type 2 DPs cannot exceed the total number of OFDM cells available for transmission of DPs:
D_DP1+D_DP2≤D_DP  [Expression 2]

where DDP1 is the number of OFDM cells occupied by Type 1 DPs, DDP2 is the number of cells occupied by Type 2 DPs. Since PLS, EAC, FIC are all mapped in the same way as Type 1 DP, they all follow “Type 1 mapping rule”. Hence, overall, Type 1 mapping always precedes Type 2 mapping.

FIG. 21, including (a) and (b), illustrates DP mapping according to an embodiment of the present invention.

FIG. 21(a) shows an addressing of OFDM cells for mapping type 1 DPs and FIG. 21(b) shows an addressing of OFDM cells for mapping for type 2 DPs.

Addressing of OFDM cells for mapping Type 1 DPs (0, . . . , DDP1-1) is defined for the active data cells of Type 1 DPs. The addressing scheme defines the order in which the cells from the TIs for each of the Type 1 DPs are allocated to the active data cells. It is also used to signal the locations of the DPs in the dynamic part of the PLS2.

Without EAC and FIC, address 0 refers to the cell immediately following the last cell carrying PLS in the last FSS. If EAC is transmitted and FIC is not in the corresponding frame, address 0 refers to the cell immediately following the last cell carrying EAC. If FIC is transmitted in the corresponding frame, address 0 refers to the cell immediately following the last cell carrying FIC. Address 0 for Type 1 DPs can be calculated considering two different cases as shown in (a). In the example in (a), PLS, EAC and FIC are assumed to be all transmitted. Extension to the cases where either or both of EAC and FIC are omitted is straightforward. If there are remaining cells in the FSS after mapping all the cells up to FIC as shown on the left side of (a).

Addressing of OFDM cells for mapping Type 2 DPs (0, . . . , DDP2-1) is defined for the active data cells of Type 2 DPs. The addressing scheme defines the order in which the cells from the TIs for each of the Type 2 DPs are allocated to the active data cells. It is also used to signal the locations of the DPs in the dynamic part of the PLS2.

Three slightly different cases are possible as shown in (b). For the first case shown on the left side of (b), cells in the last FSS are available for Type 2 DP mapping. For the second case shown in the middle, FIC occupies cells of a normal symbol, but the number of FIC cells on that symbol is not larger than CFSS. The third case, shown on the right side in (b), is the same as the second case except that the number of FIC cells mapped on that symbol exceeds CFSS.

The extension to the case where Type 1 DP(s) precede Type 2 DP(s) is straightforward since PLS, EAC and FIC follow the same “Type 1 mapping rule” as the Type 1 DP(s).

A data pipe unit (DPU) is a basic unit for allocating data cells to a DP in a frame.

A DPU is defined as a signaling unit for locating DPs in a frame. A Cell Mapper 7010 may map the cells produced by the TIs for each of the DPs. A Time interleaver 5050 outputs a series of TI-blocks and each TI-block comprises a variable number of XFECBLOCKs which is in turn composed of a set of cells. The number of cells in an XFECBLOCK, Ncells, is dependent on the FECBLOCK size, Nldpc, and the number of transmitted bits per constellation symbol. A DPU is defined as the greatest common divisor of all possible values of the number of cells in a XFECBLOCK, Ncells, supported in a given PHY profile. The length of a DPU in cells is defined as LDPU. Since each PHY profile supports different combinations of FECBLOCK size and a different number of bits per constellation symbol, LDPU is defined on a PHY profile basis.

FIG. 22 illustrates an FEC structure according to an embodiment of the present invention.

FIG. 22 illustrates an FEC structure according to an embodiment of the present invention before bit interleaving. As above mentioned, Data FEC encoder may perform the FEC encoding on the input BBF to generate FECBLOCK procedure using outer coding (BCH), and inner coding (LDPC). The illustrated FEC structure corresponds to the FECBLOCK. Also, the FECBLOCK and the FEC structure have same value corresponding to a length of LDPC codeword.

The BCH encoding is applied to each BBF (Kbch bits), and then LDPC encoding is applied to BCH-encoded BBF (Kldpc bits=Nbch bits) as illustrated in FIG. 22.

The value of Nldpc is either 64800 bits (long FECBLOCK) or 16200 bits (short FECBLOCK).

The below table 28 and table 29 show FEC encoding parameters for a long FECBLOCK and a short FECBLOCK, respectively.

TABLE 28
					BCH
					error
	LDPC				correction	Nbch-
	Rate	Nldpc	Kldpc	Kbch	capability	Kbch
	5/15	64800	21600	21408	12	192
	6/15		25920	25728
	7/15		30240	30048
	8/15		34560	34368
	9/15		38880	38688
	10/15		43200	43008
	11/15		47520	47328
	12/15		51840	51648
	13/15		56160	55968

TABLE 29
					BCH
					error
	LDPC				correction	Nbch-
	Rate	Nldpc	Kldpc	Kbch	capability	Kbch
	5/15	16200	5400	5232	12	168
	6/15		6480	6312
	7/15		7560	7392
	8/15		8640	8472
	9/15		9720	9552
	10/15		10800	10632
	11/15		11880	11712
	12/15		12960	12792
	13/15		14040	13872

The details of operations of the BCH encoding and LDPC encoding are as follows:

A 12-error correcting BCH code is used for outer encoding of the BBF. The BCH generator polynomial for short FECBLOCK and long FECBLOCK are obtained by multiplying together all polynomials.

LDPC code is used to encode the output of the outer BCH encoding. To generate a completed Bldpc (FECBLOCK), Pldpc (parity bits) is encoded systematically from each Ildpc (BCH-encoded BBF), and appended to Ildpc. The completed Bldpc (FECBLOCK) is expressed as follows.
B_ldpc=[I_ldpc P_ldpc]=[i₀,i₁, . . . ,i_Kldpc−1,p₀,p₁, . . . ,p_{Nldpc−Kldpc−1}]  [Expression 3]

The parameters for long FECBLOCK and short FECBLOCK are given in the above table 28 and 29, respectively.

The detailed procedure to calculate Nldpc−Kldpc parity bits for long FECBLOCK, is as follows:

1) Initialize the parity bits,
p₀=p₁=p₂= . . . =p_{Nldpc−Kldpc−1}=0  [Expression 4]

2) Accumulate the first information bit—i0, at parity bit addresses specified in the first row of addresses of parity check matrix. The details of addresses of parity check matrix will be described later. For example, for rate 13/15:
P₉₈₃=p₉₈₃⊕i₀ p₂₈₁₅=p₂₈₁₅⊕i₀ p₄₈₃₇=p₄₈₃₇⊕i₀ p₄₉₈₉=p₄₉₈₉⊕i₀ p₆₁₃₈=p₆₁₃₈⊕i₀ p₆₄₅₈=p₆₄₅₈⊕i₀ p₆₉₂₁=p₆₉₂₁⊕i₀ p₆₉₇₄=p₆₉₇₄⊕i₀ p₇₅₇₂=p₇₅₇₂⊕i₀ p₈₂₆₀=p₈₂₆₀⊕i₀ p₈₄₉₆=p₈₄₉₆⊕i₀  [Expression 5]

3) For the next 359 information bits, is, s=1, 2, . . . , 359 accumulate is at parity bit addresses using following expression.
{x+(s mod 360)×Q_ldpc}mod(N_ldpc−K_ldpc)  [Expression 6]

where x denotes the address of the parity bit accumulator corresponding to the first bit i0, and Qldpc is a code rate dependent constant specified in the addresses of parity check matrix. Continuing with the example, Qldpc=24 for rate 13/15, so for information bit i1, the following operations are performed:
p₁₀₀₇=p₁₀₀₇⊕i₁ p₂₈₃₉=p₂₈₃₉⊕i₁ p₄₈₆₁=p₄₈₆₁⊕i₁ p₅₀₁₃=p₅₀₁₃⊕i₁ p₆₁₆₂=p₆₁₆₂⊕i₁ p₆₄₈₂=p₆₄₈₂⊕i₁ p₆₉₄₅=p₆₉₄₅⊕i₁ p₆₉₉₈=p₆₉₉₈⊕i₁ p₇₅₉₆=p₇₅₉₆⊕i₁ p₈₂₈₄=p₈₂₈₄⊕i₁ p₈₅₂₀=p₈₅₂₀⊕i₁  [Expression 7]

4) For the 361st information bit i360, the addresses of the parity bit accumulators are given in the second row of the addresses of parity check matrix. In a similar manner the addresses of the parity bit accumulators for the following 359 information bits is, s=361, 362, . . . , 719 are obtained using the expression 6, where x denotes the address of the parity bit accumulator corresponding to the information bit i360, i.e., the entries in the second row of the addresses of parity check matrix.

5) In a similar manner, for every group of 360 new information bits, a new row from addresses of parity check matrixes used to find the addresses of the parity bit accumulators.

After all of the information bits are exhausted, the final parity bits are obtained as follows:

6) Sequentially perform the following operations starting with i=1
p_i=p_i⊕p_i-,i=1,2, . . . ,N_ldpc−K_ldpc−1  [Math Figure 8]

where final content of pi, i=0, 1, . . . Nldpc−Kldpc−1 is equal to the parity bit pi.

TABLE 30

Code Rate Qldpc


5/15      120
6/15      108
7/15      96
8/15      84
9/15      72
10/15     60
11/15     48
12/15     36
13/15     24

This LDPC encoding procedure for a short FECBLOCK is in accordance with t LDPC encoding procedure for the long FECBLOCK, except replacing the table 30 with table 31, and replacing the addresses of parity check matrix for the long FECBLOCK with the addresses of parity check matrix for the short FECBLOCK.

TABLE 31

Code Rate Qldpc


5/15      30
6/15      27
7/15      24
8/15      21
9/15      18
10/15     15
11/15     12
12/15     9
13/15     6

FIG. 23 illustrates a bit interleaving according to an embodiment of the present invention.

The outputs of the LDPC encoder are bit-interleaved, which consists of parity interleaving followed by Quasi-Cyclic Block (QCB) interleaving and inner-group interleaving.

It shows Quasi-Cyclic Block (QCB) interleaving and (b) shows inner-group interleaving.

The FECBLOCK may be parity interleaved. At the output of the parity interleaving, the LDPC codeword consists of 180 adjacent QC blocks in a long FECBLOCK and 45 adjacent QC blocks in a short FECBLOCK. Each QC block in either a long or short FECBLOCK consists of 360 bits. The parity interleaved LDPC codeword is interleaved by QCB interleaving. The unit of QCB interleaving is a QC block. The QC blocks at the output of parity interleaving are permutated by QCB interleaving as illustrated in FIG. 23, where Ncells=64800/η mod or 16200/η mod according to the FECBLOCK length. The QCB interleaving pattern is unique to each combination of modulation type and LDPC code rate.

After QCB interleaving, inner-group interleaving is performed according to modulation type and order (η mod) which is defined in the below table 32. The number of QC blocks for one inner-group, NQCB_IG, is also defined.

TABLE 32
	Modulation type	ηmod	NQCB_IG
	QAM-16	4	2
	NUC-16	4	4
	NUQ-64	6	3
	NUC-64	6	6
	NUQ-256	8	4
	NUC-256	8	8
	NUQ-1024	10	5
	NUC-1024	10	10

The inner-group interleaving process is performed with NQCB_IG QC blocks of the QCB interleaving output. Inner-group interleaving has a process of writing and reading the bits of the inner-group using 360 columns and NQCB_IG rows. In the write operation, the bits from the QCB interleaving output are written row-wise. The read operation is performed column-wise to read out m bits from each row, where m is equal to 1 for NUC and 2 for NUQ.

FIG. 24, including (a) and (b), illustrates a cell-word demultiplexing according to an embodiment of the present invention.

FIG. 24(a) shows a cell-word demultiplexing for 8 and 12 bpcu MIMO, and FIG. 24(b) shows a cell-word demultiplexing for 10 bpcu MIMO.

Each cell word (c0,1, c1,1, . . . , cη mod-1,1) of the bit interleaving output is demultiplexed into (d1,0,m, d1,1,m . . . , d1,η mod-1,m) and (d2,0,m, d2,1,m . . . , d2,η mod-1,m) as shown in (a), which describes the cell-word demultiplexing process for one XFECBLOCK.

For the 10 bpcu MIMO case using different types of NUQ for MIMO encoding, the Bit Interleaver for NUQ-1024 is re-used. Each cell word (c0,1, c1,1, . . . , c9,1) of the Bit Interleaver output is demultiplexed into (d1,0,m, d1,1,m . . . , d1,3,m) and (d2,0,m, d2,1,m . . . , d2,5,m), as shown in (b).

FIG. 25, including (a)-(c), illustrates a time interleaving according to an embodiment of the present invention.

FIG. 25(a) to FIG. 25(c) show examples of TI mode.

The time interleaver operates at the DP level. The parameters of time interleaving (TI) may be set differently for each DP.

The following parameters, which appear in part of the PLS2-STAT data, configure the TI:

DP_TI_TYPE (allowed values: 0 or 1): Represents the TI mode; ‘0’ indicates the mode with multiple TI blocks (more than one TI block) per TI group. In this case, one TI group is directly mapped to one frame (no inter-frame interleaving). ‘1’ indicates the mode with only one TI block per TI group. In this case, the TI block may be spread over more than one frame (inter-frame interleaving).

DP_TI_LENGTH: If DP_TI_TYPE=‘0’, this parameter is the number of TI blocks NTI per TI group. For DP_TI_TYPE=‘1’, this parameter is the number of frames PI spread from one TI group.

DP_NUM_BLOCK_MAX (allowed values: 0 to 1023): Represents the maximum number of XFECBLOCKs per TI group.

DP_FRAME_INTERVAL (allowed values: 1, 2, 4, 8): Represents the number of the frames HUMP between two successive frames carrying the same DP of a given PHY profile.

DP_TI_BYPASS (allowed values: 0 or 1): If time interleaving is not used for a DP, this parameter is set to ‘1’. It is set to ‘0’ if time interleaving is used.

Additionally, the parameter DP_NUM_BLOCK from the PLS2-DYN data is used to represent the number of XFECBLOCKs carried by one TI group of the DP.

When time interleaving is not used for a DP, the following TI group, time interleaving operation, and TI mode are not considered. However, the Delay Compensation block for the dynamic configuration information from the scheduler will still be required. In each DP, the XFECBLOCKs received from the SSD/MIMO encoding are grouped into TI groups. That is, each TI group is a set of an integer number of XFECBLOCKs and will contain a dynamically variable number of XFECBLOCKs. The number of XFECBLOCKs in the TI group of index n is denoted by NxBLOCK_Group(n) and is signaled as DP_NUM_BLOCK in the PLS2-DYN data. Note that NxBLOCK_Group(n) may vary from the minimum value of 0 to the maximum value NxBLOCK_Group_MAX (corresponding to DP_NUM_BLOCK_MAX) of which the largest value is 1023.

Each TI group is either mapped directly onto one frame or spread over PI frames. Each TI group is also divided into more than one TI blocks(NTI), where each TI block corresponds to one usage of time interleaver memory. The TI blocks within the TI group may contain slightly different numbers of XFECBLOCKs. If the TI group is divided into multiple TI blocks, it is directly mapped to only one frame. There are three options for time interleaving (except the extra option of skipping the time interleaving) as shown in the below table 33.

TABLE 33

Modes   Descriptions

Option- Each TI group contains one TI block and is mapped directly
1       to one frame as shown in (a). This option is signaled in the
        PLS2-STAT by DP_TI_TYPE = ‘0’ and DP_TI_
        LENGTH = ‘1’(NTI = 1).
Option- Each TI group contains one TI block and is mapped to more
2       than one frame. (b) shows an example, where one TI group
        is mapped to two frames, i.e., DP_TI_LENGTH = ‘2’
        (PI = 2) and DP_FRAME_INTERVAL (IJUMP = 2). This
        provides greater time diversity for low data-rate services.
        This option is signaled in the PLS2-STAT by DP_TI_
        TYPE = ‘1’.
Option- Each TI group is divided into multiple TI blocks and is
3       mapped directly to one frame as shown in (c). Each TI block
        may use full TI memory, so as to provide the maximum bit-
        rate for a DP. This option is signaled in the PLS2-STAT
        signaling by DP_TI_TYPE = ‘0’ and DP_TI_LENGTH =
        NTI, while PI = 1.

In each DP, the TI memory stores the input XFECBLOCKs (output XFECBLOCKs from the SSD/MIMO encoding block). Assume that input XFECBLOCKs are defined as
(d_n,s,0,0,d_n,s,0,1, . . . ,d_{n,s,0,Ncells−1},d_n,s,1,0,, . . . ,d_{n,s,1,Ncells−1}, . . . ,d_{n,s,NxBLOCK_TI(n,s)-1,0}, . . . ,d_{n,s,NxBLOCK_TI(n,s)-1,Ncells−1}),

where d_n,s,r,q is the qth cell of the rth XFECBLOCK in the sth TI block of the nth TI group and represents the outputs of SSD and MIMO encodings as follows

$d_{n,s,r,q}=\{\begin{array}{cc}{f}_{n,s,r,q},& \mathrm{the}\mathrm{output}\mathrm{of}\mathrm{SSD}\dots \mathrm{encoding}\\ g_{n,s,r,q},& \mathrm{the}\mathrm{output}\mathrm{of}\mathrm{MIMO}\dots \mathrm{encoding}\end{array}.$

In addition, assume that output XFECBLOCKs from the time interleaver 5050 are defined as
(h_n,s,0,h_n,s,1, . . . ,h_n,s,i, . . . ,h_{n,s,NxBLOCK_TI(n,s)×Ncells−1}),

where h_n,s,i is the ith output cell (for i=0, . . . , N_{xBLOCK_TI}(n,s)×N_cells−1) in the sth TI block of the nth TI group.

Typically, the time interleaver will also act as a buffer for DP data prior to the process of frame building. This is achieved by means of two memory banks for each DP. The first TI-block is written to the first bank. The second TI-block is written to the second bank while the first bank is being read from and so on.

The TI is a twisted row-column block interleaver. For the sth TI block of the nth TI group, the number of rows N_r of a TI memory is equal to the number of cells N_cells, i.e., N_r=N_cells while the number of columns N_c is equal to the number N_{xBLOCK_TI}(n,s).

FIG. 26, including (a) and (b), illustrates the basic operation of a twisted row-column block interleaver according to an embodiment of the present invention.

FIG. 26(a) shows a writing operation in the time interleaver and FIG. 26(b) shows a reading operation in the time interleaver. The first XFECBLOCK is written column-wise into the first column of the TI memory, and the second XFECBLOCK is written into the next column, and so on as shown in FIG. 26(a). Then, in the interleaving array, cells are read out diagonal-wise. During diagonal-wise reading from the first row (rightwards along the row beginning with the left-most column) to the last row, N_r cells are read out as shown in (b). In detail, assuming z_n,s,i(i=0, . . . , N_rN_c) as the TI memory cell position to be read sequentially, the reading process in such an interleaving array is performed by calculating the row index R_n,s,i, the column index C_n,s,i, and the associated twisting parameter T_n,s,i as follows expression.

$\begin{array}{cc}\mathrm{GENERATE}\left(R_{n,s,i},C_{n,s,i}\right)=\left\{R_{n,s,i}=\mathrm{mod}\left(i,N_r\right),T_{n,s,i}=\mathrm{mod}\left(S_{\mathrm{shift}}\times R_{n,s,i},N_c\right),C_{n,s,i}=\mathrm{mod}\left(T_{n,s,i}+\lfloor \frac{i}{N_r}\rfloor ,N_c\right)\right\}& \left[\mathrm{Expression}9\right]\end{array}$

where S_shift is a common shift value for the diagonal-wise reading process regardless of N_{xBLOCK_TI}(n,s), and it is determined by N_{xBLOCK_TI_MAX} given in the PLS2-STAT as follows expression.

$\begin{array}{cc}\mathrm{for}\{\begin{array}{cc}{N}_{\mathrm{xBLOCK}\_\mathrm{TI}\_\mathrm{MAX}}^{\prime }=N_{\mathrm{xBLOCK}\_\mathrm{TI}\_\mathrm{MAX}}+1,& \mathrm{if}{N}_{\mathrm{xBLOCK}\_\mathrm{TI}\_\mathrm{MAX}}\mathrm{mod}2=0\\ N_{\mathrm{xBLOCK}\mathrm{TI}\mathrm{MAX}}^{\prime }=N_{\mathrm{xBLOCK}\mathrm{TI}\mathrm{MAX}}& \mathrm{if}{N}_{\mathrm{xBLOCK}\mathrm{TI}\mathrm{MAX}}\mathrm{mod}2=1\end{array},& \left[\mathrm{Expression}10\right]\end{array}$ $S_{\mathrm{shift}}=\frac{N_{\mathrm{xBLOCK}\_\mathrm{TI}\_\mathrm{MAX}}^{\prime }-1}{2}$

As a result, the cell positions to be read are calculated by a coordinate as z_n,s,i=N_rC_n,s,i+R_n,s,i.

FIG. 27 illustrates an operation of a twisted row-column block interleaver according to another embodiment of the present invention.

More specifically, FIG. 27 illustrates the interleaving array in the TI memory for each TI group, including virtual XFECBLOCKs when N_{xBLOCK_TI}(0,0)=3, N_{xBLOCK_TI}(1,0)=6, N_{xBLOCK_TI}(2,0)=5.

The variable number N_{xBLOCK_TI}(n,s)=N_r will be less than or equal to N′_{xBLOCK_TI_MAX}. Thus, in order to achieve a single-memory deinterleaving at the receiver side, regardless of N_{xBLOCK_TI}(n,s), the interleaving array for use in a twisted row-column block interleaver is set to the size of N_r×N_c=N_cells×N_{xBLOCK_TI_MAX} by inserting the virtual XFECBLOCKs into the TI memory and the reading process is accomplished as follow expression.

(Expression 11)


p = 0;
for i = 0;i < NcellsN′xBLOCK_TI_MAX;i = i + 1
{GENERATE (Rn,s,i, Cn,s,i);
Vi = NrCn,s,j + Rn,s,j
if Vi < NcellsNXBLOCK_TI(n,s)
{
Zn,s,p = Vi; p = p + 1;
}
}

The number of TI groups is set to 3. The option of time interleaver is signaled in the PLS2-STAT data by DP_TI_TYPE=‘0’, DP_FRAME_INTERVAL=‘1’, and DP_TI_LENGTH=‘1’, i.e., NTI=1, IJUMP=1, and PI=1. The number of XFECBLOCKs, each of which has Ncells=30 cells, per TI group is signaled in the PLS2-DYN data by NxBLOCK_TI(0,0)=3, NxBLOCK_TI(1,0)=6, and NxBLOCK_TI(2,0)=5, respectively. The maximum number of XFECBLOCK is signaled in the PLS2-STAT data by NxBLOCK_Group_MAX, which leads to └N_{xBLOCK_Group_MAX}/N_TI┘=N_{xBLOCK_TI_MAX}=6.

FIG. 28 illustrates a diagonal-wise reading pattern of a twisted row-column block interleaver according to an embodiment of the present invention.

More specifically FIG. 28 shows a diagonal-wise reading pattern from each interleaving array with parameters of N′_{xBLOCK_TI_MAX}=7 and Sshift=(7−1)/2=3. Note that in the reading process shown as pseudocode above, if V_i≥N_cellsN_{xBLOCK_TI}(n,s), the value of Vi is skipped and the next calculated value of Vi is used.

FIG. 29 illustrates interleaved XFECBLOCKs from each interleaving array according to an embodiment of the present invention.

FIG. 29 illustrates the interleaved XFECBLOCKs from each interleaving array with parameters of N′_{xBLOCK_TI_MAX}=7 and Sshift=3.

The frequency interleaving process according to one embodiment of the present invention will hereinafter be described.

The frequency interleaver 7020 according to one embodiment is configured to apply different interleaving sequences to respective cells corresponding to each OFDM symbol so as to improve the frequency diversity performance in the OFDM symbol structure composed of a plurality of cells.

In the present invention, the above-mentioned frequency interleaving method may be referred to as ‘random frequency interleaving’ or ‘random FI’, and may also be changed according to designer's intention.

The above-mentioned broadcast signal transmission apparatus or the frequency interleaver 7020 contained in the broadcast signal transmission apparatus may apply different interleaving sequences either to constituent symbols (i.e., respective symbols) of at least one frame or to respective cells corresponding to two paired symbols, and may perform frequency interleaving, thereby obtaining the frequency diversity.

The at least one symbol may be converted into at least one OFDM symbols in a subsequent modulation process. That is, the at least one symbol may be converted into each OFDM symbol or two paired OFDM symbols (i.e., pair-wise OFDM symbol or each OFDM symbol pair).

The frequency interleaver according to one embodiment may perform frequency interleaving of cells corresponding to OFDM symbols entered using the frequency interleaving address generated on the basis of the main interleaving sequence (or basic interleaving sequence) and the symbol offset.

Hereinafter, a description will be given of MIMO processing (or a MIMO encoding method or decoding method) according to an embodiment of the present invention.

A next generation broadcast system according to an embodiment of the present invention may use a MIMO scheme to deliver more data. In particular, the next generation broadcast system according to the present embodiment may overcome a correlated channel environment in which channel capacity is decreased and system operation is disturbed using dual-polar MIMO in which a signal is transmitted using a vertical/horizontal polarity of an antenna. However, even when dual-polar MIMO is used, two components may have different power ratios according to difference in radio wave propagation characteristic of a vertical/horizontal polarity. In other words, a power imbalance state may occur. In this state, power is different between a signal output from an antenna of a horizontal polarity and a signal output from an antenna of a vertical polarity. To overcome this power imbalance environment, the MIMO scheme according to the present embodiment may include a PH-eSM scheme (PH-eSM PI) that requires less complexity of a receiver, a full-rate full-diversity (FRFD) scheme (FRFD PH-eSM PI) that can exhibit higher performance even though a design of the receiver is complex, and a scheme created by combining each of the above schemes with a non-uniform constellation (NUC). The PH-eSM scheme refers to a scheme of rotating angles of signals input to a MIMO decoder to overcome complexity of the receiver and power imbalance. The FRFD scheme is a scheme of acquiring additional diversity gain in the frequency domain by adding additional complexity.

When MIMO is combined with an NUC, the NUC uses a grid value optimized for each SNR (or code rate of FEC or combination of modulation and code rate (MODCOD)). Thus, when MIMO encoding is performed, a transmitting end needs to use a parameter “a” of MIMO encoding having an optimized value for each SNR.

Hereinafter, a scheme will be proposed to optimize the MIMO parameter “a” according to SNR or MODCOD in a MIMO system to which an NUC (1D-NUC or 2D-NUC) is applied. In addition, a description will be given of a method of reducing complexity of the receiver using a decoder suitable for an NUC (for example, a sphere decoder (SD) and a complex-sphere decoder).

FIG. 30, including (a) and (b), shows equations that express a matrix according to the MIMO scheme according to an embodiment of the present invention.

Specifically, FIG. 30(a) is an equation that expresses the PH-eSM scheme or a MIMO matrix according to the PH-eSM scheme, and FIG. 30(b) is an equation that expresses the FRFD scheme or a MIMO matrix according to the FRFD scheme.

As described in the foregoing, the PH-eSM scheme is a scheme that reduces complexity according to MIMO decoding. A MIMO processor may generate signals Xi and X₂ to be transmitted using two transmitted symbols (input symbols or a symbol pair) output from a constellation mapper, for example, QAM symbols S₁ and S₂. In an OFDM transmission/reception system, the signals X₁ and X₂ may be transmitted on the same frequency carrier f₁ (X₁(f₁), X₂(f₁)). Here, the signal X₁ may be transmitted through an antenna 1 (TX1), and the signal X₂ may be transmitted through an antenna 2 (TX2). According to this scheme, even when power imbalance is present between two transmit antennas, it is possible to implement efficient signal transmission that minimizes loss.

In a left part of the figure, a value X refers to an output of MIMO processing of a transmitter, S refers to an input signal, and P refers to a MIMO matrix. P may be referred to as a precoding matrix.

The precoding matrix may include a rotation matrix for combining constellations of input symbols by rotating angles of the input symbols, and a phase-hopping matrix for configuring phase rotation of symbols transmitted through the antenna 2 (TX2).

“a” in the rotation matrix may be referred to as a MIMO parameter. When input symbols correspond to QAM symbols, a value thereof may be determined according to QAM order. When the receiver receives the signals X₁ and X₂ and independently decodes the signals X₁ and X₂, that is, when the receiver decodes the symbols S₁ and S₂ only using the signal X₁ or decodes the symbols S₁ and S₂ only using the signal X₂, a value of “a” is optimized based on Euclidean distance and Hamming distance. Thus, the receiver may obtain low bit error rate (BER) performance. The receiver according to the present embodiment may perform ML and sub-ML (sphere) decoding, etc. using the matrix shown in the figure to perform a reverse operation of MIMO processing at a transmitting side.

A parameter 0 of the phase-hopping matrix indicates a phase rotation angle, and a value thereof is determined according to the equation shown in the figure.

The FRFD scheme is a scheme of obtaining high performance by performing more complex decoding in the receiver.

As illustrated in the figure, the MIMO processor on the transmitting side may receive four input symbols S₁, S₂, S₃, and S₄ and output transmitted signals X₁ and X₂ using the matrix shown in the figure. In this case, the same value as a parameter according to the PH-eSM scheme may be used as the MIMO parameter a.

In addition, in the FRFD scheme, the output signals X₁ and X₂ may be transmitted on two frequency carriers f₁ and f₂ (X₁(f₁), X₂(f₁), X₁(f₂), X₂(f₂)). Therefore, the signal X₁ is transmitted through two frequency carriers of the antenna 1 (TX1), and the signal X₂ is transmitted through two frequency carriers of the antenna 2 (TX2). Thus, this scheme has an advantage of effectively coping with a case in which power imbalance occurs between carriers in addition to power imbalance between antennas.

FIG. 31, including (a) and (b), shows the MIMO parameter “a” when an NUC is applied according to an embodiment of the present invention.

The above-described MIMO parameter “a” may have a different value according to uniform QAM and 1D-NUC modulation value of an input symbol.

FIG. 31(a) shows examples of a value of the MIMO parameter “a” according to QAM combinations, and FIG. 31(b) shows a process of determining the value of the MIMO parameter “a” according to 1D-NUC modulation and the value a according to each modulation combination.

As shown in FIG. 31(a), the value of the MIMO parameter “a” may be set according to QAM combinations, that is, QPSK, 16 QAM, 64 QAM, and 256 QAM.

In addition, as shown in FIG. 31(b), in a 1D-NUC modulation combination, the value of the MIMO parameter “a” may be calculated by defining a PAM grid of a used I or Q component and using a largest value (Pm) and a second largest value (Pm−1) of the grid. Therefore, the receiver may separately decode a signal received through one antenna by applying the value a.

Further, as shown in the figure, the value a may be calculated using a sub-constellation separation factor b. The factor b refers to a parameter for adjusting an interval between sub-constellations present in a MIMO-processed signal. In an example of the present invention, the factor b has a value of

$\frac{\sqrt{2}}{2},$
The value b refers to a value obtained by adjusting a Euclidean distance and a Hamming distance based on a point having largest power and an adjacent point thereof in constellations.

However, in 1D-NUC, a grid value optimized for each code rate of 1-EC or SNR is used, and thus the MIMO parameter “a” needs to have an optimized value for each SNR.

Therefore, the present invention proposes a method of calculating a value of the MIMO parameter “a” optimized for each SNR. To apply an NUC to the above-described two MIMO schemes for each SNR (or FEC code rate), the following two factors need to be considered.

First, an NUC, which is most suitable for MIMO, needs to be found to obtain shaping gain. Second, a value of the MIMO parameter “a” needs to be calculated in an NUC optimized for each SNR.

However, when an NUC suitable for each SNR and the MIMO parameter “a” are simultaneously calculated through capacity analysis of BICM in a MIMO environment, the calculation may be restricted due to operation quantity. Therefore, the present invention proposes the following process.

First, an NUC suitable for MIMO is determined by comparing performances of NUCs optimized for SISO on a non-power imbalanced MIMO channel for each SNR (or FEC code rate). When a BICM capacity can be analyzed on a MIMO channel, it is possible to obtain an optimized NUC for MIMO which provides a maximum capacity at a certain SNR by analyzing capacities according to SNRs.

However, when operation quantity is too large to perform actual calculation, an NUC for MIMO may be determined by comparing performances using an optimized NUC set (NUC for each SNR) in SISO. For example, an NUC for MIMO at a code rate of 5/15 of 12 bpcu (1D-64NUC+1D-64NUC) may be determined to be 1D-64NUC corresponding to an SISO code rate of 5/15.

Thereafter, a MIMO parameter “a” optimized for each SNR may be calculated by analyzing capacities in a power imbalanced MIMO channel environment based on an NUC determined through the above-described process.

FIG. 32, including (a) and (b), shows an equation that expresses a MIMO scheme in consideration of power imbalance and a correlated channel environment and an equation that expresses a capacity of BICM according to an embodiment of the present invention.

Specifically, FIG. 32(a) shows an equation that expresses a MIMO scheme in consideration of power imbalance and a correlated channel environment, and FIG. 32(b) shows an equation that expresses a capacity of BICM.

As shown in FIG. 32(a), according to an embodiment of the present invention, a power imbalance (PI) factor α and an active white Gaussian noise (AWGN) value are applied to the above-described MIMO matrix in order to consider a correlated channel environment and a power imbalance channel environment which can be generated when the MIMO scheme is used. The factor α according to the present embodiment may have values as below according to magnitudes of PI.

PI 9 dB: 0.354817, PI 6 dB: 0.501187, PI 3 dB: 0.70711

In addition, as shown in FIG. 32(b), a BICM capacity may be determined on the assumption of AWGN and individually identically distributed (IID) input.

In other words, an algorithm for obtaining a value of the parameter a proposed in the present invention is as below.

First step: An initial NUC of an SNR or a MIMO FEC code rate (CR) having optimum performance is obtained by comparing BER performance using an NUC corresponding to an FEC code rate (5/15, 6/15, 7/15, . . . , 13/15) of SISO. For example, an NUC of a MIMO FEC CR of 6/15 may have an NUC corresponding to an SISO FEC CR of 5/15.

Second step: A MIMO parameter “a” having optimum performance may be calculated by analyzing BICM performance as described above with reference to FIG. 32 with respect to the NUC obtained in the first step. In this case, a value of the parameter a may be calculated based on a power imbalance and a correlated channel environment described above with reference to FIG. 32.

FIG. 33, including (a) and (b), shows an equation that expresses a condition of a MIMO decoder for an NUC.

A receiver according to an embodiment of the present invention may perform a reverse operation of MIMO processing on the transmitting side. In addition, this figure illustrates a MIMO decoder for reducing complexity in a MIMO system to which an NUC is applied.

An SD refers to a MIMO decoder having reduced complexity while having maximum-likelihood (ML) performance Therefore, the MIMO decoder according to the present embodiment may find a constellation point free of noise within a radius R of a hyper-sphere around a received signal r. The equation shown in FIG. 33(a) expresses ML according to a relation among the received signal r, the constellation point free of noise, and the radius R of the hyper-sphere, and FIG. 33(b) illustrates constellation points free of noise within the value of R of the hyper-sphere around the received signal. In the equation, HS denotes the constellation point free of noise.

Most SDs are based on a real-valued SD which is based on a constellation based on QAM or PSK. However, these SDs are not suitable for application to an NUC such as 2D-NUC. Therefore, an embodiment of the present invention may be based on a complex SD suitable for 2D-NUC. The complex SD suitable for 2D-NUC is advantageous in optimizing hardware execution. Thus, a value of an initial radius R may be selected according to change in noise for each antenna, and complexity of an algorithm may be based on a noise level and a channel condition. In other words, according to an embodiment of the present invention, a broadcast signal transmission/reception system may use a value of R according to a constellation of each MODCOD to optimize MIMO performance

FIG. 34 shows MIMO constellations and MIMO parameters “a” proposed for 8 bpcu, 12 bpcu, and 16 bpcu according to an embodiment of the present invention.

A leftmost column of a table shown in the figure indicates code rates, and an upper row of the table indicates values of bpcu.

When a constellation according to an embodiment of the present invention is used, it is possible to obtain shaping gain optimized for each code rate (MODCOD). In addition, a MIMO parameter “a” according to the present embodiment may optimize MIMO performance when a constellation optimized for each MODCOD is applied. Further, optimum performance for each MODCOD may be acquired by applying a constellation and a MIMO parameter “a” according to the present embodiment.

FIG. 35 shows MIMO constellations and MIMO parameters “a” proposed for 8 bpcu, 12 bpcu, and 16 bpcu according to another embodiment of the present invention.

This figure shows a constellation that optimizes performance while maintaining a baseline of SISO for a constellation acquired through the above-described method for 8 bpcu, 12 bpcu, and 16 bpcu.

A leftmost column of a table shown in the figure indicates code rates, and an upper row of the table indicates values of bpcu. A MIMO parameter “a” is a value that can optimize MIMO performance when a constellation optimized for each MODCOD is applied.

When the constellation according to the embodiment of the present invention is used, it is possible to obtain optimized shaping gain while maintaining a baseline of SISO for each code rate (MODCOD) as much as possible. Therefore, additional configurations for MIMO may be reduced in the transmitter and the receiver, and thus complexity of a transmission/reception configuration may be reduced. In addition, it is possible to obtain optimum performance for each MODCOD by applying the constellation and the MIMO parameter “a” according to the present embodiment.

FIG. 36 shows MIMO constellations and MIMO parameters “a” proposed for 20 bpcu and 24 bpcu according to an embodiment of the present invention.

The present embodiment is different from the embodiments described above with reference to FIGS. 34 and 35 in values of bpcu, and details thereof are the same. Thus, a description of the details will be omitted.

FIG. 37 shows a type of a precoding matrix of the PH-eSM scheme according to an embodiment of the present invention.

An equation shown at the top of the figure expresses a MIMO matrix of the above-described PH-eSM scheme, and four types of precoding matrices in the equation shown at the top of the figure are shown at the bottom of the figure.

As described in the foregoing, the MIMO matrix of the PH-eSM scheme may include a precoding matrix (P), and the precoding matrix may include a rotation matrix and a phase rotation matrix. The rotation matrix may include the above-described MIMO parameter “a”, and the rotation matrix may be expressed in the four types as shown at the bottom of the figure.

The four types of precoding matrices shown at the bottom of the figure are different from one another only in terms of expression and are substantially the same. That is, the MIMO parameter “a” may be expressed using a rotation angle θ.

In the precoding matrix of Type 4, a value of the matrix may be changed according to a value of n without fixing a rotation angle of a rotation matrix used for precoding. In this case, the value of n may change according to an OFDM carrier index or change according to an OFDM symbol index. The receiver may decode the value of n in synchronization with the transmitter. Therefore, when the precoding matrix of Type 4 is used, there is an advantage of obtaining average performance of several rotation angles without being affected by MIMO encoding performance which depends on a particular rotation angle of a rotation matrix.

In this case, a value of “a” and a value of a rotation angle may be determined based on a constellation value. When the signals X₁ and X₂ respectively transmitted through the antenna 1 (TX1) and the antenna 2 (TX2) are independently decoded, that is, when the symbols S₁ and S₂ are decoded only using the signal X₁ or only using the signal X₂, a value of “a” is optimized based on Euclidean distance and Hamming distance. Thus, the receiver may obtain low BER performance

The receiver may perform ML and sub-ML (sphere) decoding, etc. using the matrix shown in the figure on signals MIMO-processed using the matrix, thereby performing a reverse operation of MIMO processing on the transmitting side.

FIG. 38 shows a MIMO matrix of the FRFD scheme according to an embodiment of the present invention.

As described in the foregoing, the FRFD scheme is a scheme of obtaining high performance by performing more complex decoding in the receiver. Details are the same as the above description with reference to FIG. 30. Four types of matrices shown in the figure are different from one another only in terms of expression and are substantially the same.

FIG. 39 shows signaling information for signaling a MIMO matrix according to an embodiment of the present invention.

The present invention proposes a method of transmitting a value of a MIMO parameter “a” using a signaling field (or signaling information) for signaling the value to the receiver. The signaling field may be expressed by MIMO encoding_parameter, and have a size of 4 bits. In addition, a bit resolution of a MIMO parameter may vary according to size of the signaling field.

The signaling field according to the present embodiment may be included in PLS1 or PLS2 described above. In addition, PLS1 and PLS2 according to the present embodiment may be referred to as L1 signaling information. The receiver according to the present embodiment may receive the above-described signaling information to perform flexible decoding.

A signaling field indicating a MIMO parameter (or MIMO encoding parameter) of a matrix corresponding to Type 2 is shown on the left side of the figure, and a signaling field indicating a MIMO parameter (or MIMO encoding parameter) of a matrix corresponding to Type 3 is shown on the right side of the figure.

Each of the MIMO parameter of the matrix corresponding to Type 2 and the MIMO parameter of the matrix corresponding to Type 3 may be defined as “a” or a rotation angle. A relation between the two parameters is shown at the bottom of the figure.

Therefore, in Type 2, “a” corresponds to 0 when a value of MIMO encoding_parameter corresponds to 0, and “a” corresponds to 1 when a value of MIMO encoding_parameter corresponds to 9. In a similar manner, in Type 3, a rotation angle corresponds to 0° when a value of MIMO encoding_parameter corresponds to 0, and a rotation angle corresponds to 45° when a value of MIMO encoding_parameter corresponds to 9.

In addition, a bit resolution of a MIMO parameter according to a value of MIMO encoding_parameter may change. Further, in Type 3, a resolution between MIMO parameters determined according to a value of MIMO encoding_parameter based on 4 bits may correspond to 5 degrees.

MIMO parameters of matrices corresponding to Type 1 and Type 4 according to an embodiment of the present invention may have similar values to those of MIMO parameters of matrices corresponding to Type 2 and Type 3.

FIG. 40 shows MIMO constellations and MIMO parameters proposed for 4 bpcu and 8 bpcu according to another embodiment of the present invention.

FIG. 40 shows optimized MIMO parameters when an ML decoder is applied to 4 bpcu (QPSK+QPSK) and 8 bpcu (16NUC+16NUC).

FIG. 40 shows rotation angles as another embodiment of the MIMO parameters described above with reference to FIGS. 34 to 36. In addition, a rotation angle may be expressed using “a” according to an equation shown at the right top of the table.

A leftmost column of the table shown in the figure indicates code rates, and an upper row of the table indicates values of bpcu.

When the receiver performs ML decoding, optimum MIMO performance may be obtained by applying the MIMO parameters shown in the figure.

A MIMO parameter rotation angle according to an embodiment of the present invention may optimize MIMO performance when a constellation optimized for each MODCOD is applied. In addition, optimum performance may be obtained for each MODCOD by applying a constellation and a MIMO parameter rotation angle according to the present embodiment.

FIG. 41 shows MIMO constellations and MIMO parameters proposed for 12 bpcu, 16 bpcu and 20 bpcu according to another embodiment of the present invention.

FIG. 41 shows optimized MIMO parameters when a linear receiver is applied to 12 bpcu (64NUC+64NUC), 16 bpcu (256NUC+256NUC), and 20 bpcu (1024NUC+1024NUC).

The linear receiver according to the present embodiment may include a MIMO demapper using an equalizer based on an MMSE filter. In this case, optimum MIMO performance may be obtained when the MIMO parameters shown in the figure are used. When the MIMO parameters and a linear type-reception configuration are used, there is an advantage of reducing complexity of the receiver while minimizing performance deterioration.

A MIMO parameter rotation angle according to an embodiment of the present invention may optimize MIMO performance when a constellation optimized for each MODCOD is applied. In addition, optimum performance may be obtained for each MODCOD by applying a constellation and a MIMO parameter rotation angle according to the present embodiment.

FIG. 42 shows MIMO constellations and MIMO parameters proposed for 24 bpcu according to another embodiment of the present invention.

FIG. 42 shows optimized MIMO parameters when a linear receiver is applied to 24 bpcu (2048NUC+2048NUC). The present embodiment is different from the embodiment described above with reference to FIG. 41 in values of bpcu, and details thereof are the same. Thus, the details will be omitted.

FIG. 43 is a flowchart illustrating a method for receiving broadcast signals according to an embodiment of the present invention.

The apparatus for receiving broadcast signals according to an embodiment of the present invention may perform a reverse process of transmitting broadcast signals which is described in FIG. 1 to FIG. 8, FIG. 10 to FIG. 57.

The apparatus for receiving broadcast signals according to an embodiment of the present invention or a receiver can receive broadcast signals (S43000).

Then the apparatus for receiving broadcast signals according to an embodiment of the present invention or a synchronization & demodulation module in the apparatus for receiving broadcast signals can demodulate the received broadcast signals by an OFDM (Orthogonal Frequency Division Multiplexing) scheme (S43100). Details are as described in FIG. 9.

The apparatus for receiving broadcast signals according to an embodiment of the present invention or the frame parsing module can parse at least one signal frame from the frequency deinterleaved broadcast signals (S43200). The detailed process of parsing is as described in FIG. 9.

Then the apparatus for receiving broadcast signals according to an embodiment of the present invention or the demapping & decoding module in the apparatus for receiving broadcast signals can MIMO process service data in the parsed signal frame (S43300). More specifically, the MIMO processing of an apparatus for transmitting broadcast signals is performed on a pair of symbols of the mapped service data based on a rotation matrix with a rotation angle. The value of the rotation angle depends on a modulation order and code rate of the service data and the modulation order is one of QPSK modulation and NUC (non-uniform constellation). The details of MIMO processing are as described in FIG. 30 to FIG. 42. The apparatus for receiving broadcast signals according to an embodiment of the present invention can perfume MIMO processing as a reverse process of the apparatus for transmitting.

Then, the apparatus for receiving broadcast signals according to an embodiment of the present invention or the demapping & decoding module can de-map the MIMO processed service data (S43400) and decode the demapped service data (S43500). The detailed process of the two steps is as described in FIG. 9.

It will be appreciated by those skilled in the art that various modifications and variations can be made in the present invention without departing from the spirit or scope of the inventions. Thus, it is intended that the present invention covers the modifications and variations of this invention provided they come within the scope of the appended claims and their equivalents.

Both apparatus and method inventions are mentioned in this specification and descriptions of both of the apparatus and method inventions may be complementarily applicable to each other.

A module, a unit or a block according to embodiments of the present invention is a processor/hardware executing a sequence of instructions stored in a memory (or storage unit). The steps or the methods in the above described embodiments can be operated in/by hardwares/processors. In addition, the method of the present invention may be implemented as a code that may be written on a processor readable recording medium and thus, read by the processors provided in the apparatus according to embodiments of the present invention.

Claims

1. A method for processing broadcast signals, the method comprising:

receiving the broadcast signals carrying service data;

demodulating the broadcast signals by an orthogonal frequency division multiplexing (OFDM) scheme;

multiple input multiple output (MIMO) processing data in the broadcast signals based on a rotation value,

the rotation value has zero (0) degree for a modulation order corresponding to equal to or greater than 64 quadrature amplitude modulation (QAM) and a code rate corresponding to at least one of 2/15, 3/15, 4/15, 5/15, 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15,

the rotation value has zero (0) degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 2/15, 3/15, 4/15 or 5/15,

the rotation value has non-zero degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15; and

decoding the data.

2. The method of claim 1,

wherein the data in the MIMO processing is rotated based on a matrix having the rotation value.

3. A device for processing broadcast signals, the device comprising:

a tuner configured to receive the broadcast signals carrying service data;

a demodulator configured to demodulate the broadcast signals by an orthogonal frequency division multiplexing (OFDM) scheme;

a multiple input multiple output (MIMO) processor configured to MIMO process data in the broadcast signals based on a rotation value,

the rotation value has zero (0) degree for a modulation order corresponding to equal to or greater than 64 quadrature amplitude modulation (QAM) and a code rate corresponding to at least one of 2/15, 3/15, 4/15, 5/15, 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15,

the rotation value has zero (0) degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 2/15, 3/15, 4/15 or 5/15,

the rotation value has non-zero degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15; and

a decoder configured to decode the data.

4. The device of claim 3, wherein the data, by the MIMO processor, is rotated based on a matrix having the rotation value.

5. A method for transmitting broadcast signals, the method comprising:

encoding data;

multiple input multiple output (MIMO) processing the data based on a rotation value,

the rotation value has zero (0) degree for a modulation order corresponding to equal to or greater than 64 quadrature amplitude modulation (QAM) and a code rate corresponding to at least one of 2/15, 3/15, 4/15, 5/15, 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15,

the rotation value has zero (0) degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 2/15, 3/15, 4/15 or 5/15,

the rotation value has non-zero degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15;

modulating broadcast signals including the data by an orthogonal frequency division multiplexing (OFDM) scheme; and

transmitting the broadcast signals.

6. An apparatus for transmitting broadcast signals, the apparatus comprising:

an encoder configured to encode data;

a multiple input multiple output (MIMO) processor configured to MIMO process the data based on a rotation value,

the rotation value has zero (0) degree for a modulation order corresponding to equal to or greater than 64 quadrature amplitude modulation (QAM) and a code rate corresponding to at least one of 2/15, 3/15, 4/15, 5/15, 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15,

the rotation value has zero (0) degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 2/15, 3/15, 4/15 or 5/15,

the rotation value has non-zero degree for a modulation order corresponding to quadrature phase Shift Keying (qpsk) and a code rate corresponding to at least one of 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 or 13/15;

a modulator configured to modulate broadcast signals including the data by an orthogonal frequency division multiplexing (OFDM) scheme; and

a transmitter configured to transmit the broadcast signals.

7. An apparatus for receiving broadcast signals, the apparatus comprising:

a receiver to receive the broadcast signals carrying service data that are processed by a multiple-input multiple-output (MIMO) operation;

a demodulator to demodulate the broadcast signals by an orthogonal frequency division multiplexing (OFDM) scheme,

wherein the MIMO operation is based on rotation matrices,

wherein values of rotation angles of the rotation matrices depend on modulation orders and code rates of the service data;

a demapper to demap data in the broadcast signals; and

a decoder to decode the data based on the code rates,

wherein when a modulation order is quadrature phase Shift Keying (qpsk) and the code rates are 2/15, 3/15, 4/15, and 5/15, the values of the rotation angles are zero (0) degrees,

wherein when a modulation order is qpsk and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees.

8. The apparatus of claim 7,

wherein when a modulation order is a Non-Uniform-Constellation (NUC) 64 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees,

wherein when a modulation order is a NUC-256 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees,

wherein when a modulation order is a NUC-1024 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees.

9. A method for receiving broadcast signals, the method comprising:

receiving the broadcast signals carrying service data that are processed by a multiple-input multiple-output (MIMO) operation;

demodulating the broadcast signals by an orthogonal frequency division multiplexing (OFDM) scheme,

wherein the MIMO operation is based on rotation matrices,

wherein values of rotation angles of the rotation matrices depend on modulation orders and code rates of the service data;

demapping data in the broadcast signals; and

decoding the data based on the code rates,

wherein when a modulation order is quadrature phase Shift Keying (qpsk) and the code rates are 2/15, 3/15, 4/15, and 5/15, the values of the rotation angles are zero (0) degrees,

wherein when a modulation order is qpsk and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees.

10. The method of claim 9,

wherein when a modulation order is a Non-Uniform-Constellation (NUC) 64 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees,

wherein when a modulation order is a NUC-256 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees,

wherein when a modulation order is a NUC-1024 and the code rates are 6/15, 7/15, 8/15, 9/15, 10/15, 11/15, 12/15 and 13/15, the values of the rotation angles are non-zero (0) degrees.